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Can I take metformin if I have kidney problems? Jack Merendino. However, in people who have kidney disease, usually with an elevated creatinine level, metformin increases the risk of lactic acidosis, so the drug should not be used. In other situations where the kidney function may be reduced, such as in people who have heart failure.
doi: 10.1503/cmaj.045292
PMID: 16129871
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Metformin has been used for over 40 years for patients with type 2 diabetes mellitus. With over 40 million patient-years of use as of 1999, there is now evidence that the drug decreases the risk of morbidity and death when used to treat type 2 diabetes. However, concern remains over the possible side effect of lactic acidosis, a condition with a mortality of up to 50%. Because of this concern, contraindications to the use of metformin have been suggested. In particular, the following 3 are listed in the Compendium of Pharmaceuticals and Specialties:4
These specific contraindications cause consternation for clinicians who may wish to prescribe metformin but do not want to put their patients at risk of lactic acidosis or expose themselves to potential legal problems. However, we believe the evidence shows that the benefits of metformin use in patients with contraindications outweigh the risks. In 2002 Calabrese and associates reported that 62% of 204 hospital patients taking metformin had at least 1 contraindication or precautionary condition (renal impairment, congestive heart failure, age ≥ 80 years, exposure to contrast media, hepatic disease, excessive alcohol intake, surgery). In particular, 14% of the patients taking metformin had an elevated creatinine level and 10% were older than 80. These findings led Calabrese and associates to state that “many patients are treated with metformin despite having clinical conditions that place them at risk for developing lactic acidosis. To minimize this risk, it is essential that prescribers develop a better understanding of the prescribing guidelines for metformin.” Interestingly, despite the presence of these risks, lactic acidosis did not develop in any of the patients in their study. Although these authors' intentions were good, they and other clinicians who avoid metformin use in these patients, in our opinion, have not made an evidence-based decision. In a study from Germany that examined the prescribing of metformin outside of its clinical recommendations, 73% of 308 patients had at least 1 contraindication to the drug; despite this, no cases of lactic acidosis were found. In a study from Scotland involving 1847 patients taking metformin, the drug was prescribed outside of guidelines in 24.5% of cases; despite this, only 1 case of lactic acidosis occurred over a 30-month period, and its cause was secondary to cardiac failure. An estimate of the absolute benefit of metformin use in terms of macrovascular end points can be taken from the results of the UK Prospective Diabetes Study (UKPDS). In this trial, absolute reductions were observed over 10 years in the number of diabetes-related deaths (by 5%), all-cause mortality (by 7%), myocardial infarction (by 6%) and stroke (by 3%) among patients with newly diagnosed type 2 diabetes who were given metformin compared with those following dietary advice. In the same study, the use of insulin and glyburide did not lead to any reductions in the risk of macrovascular disease compared with dietary advice alone. So far, the only other drug with any evidence of reducing macrovascular complications in type 2 diabetes is acarbose. The conclusions from the UKPDS and from a recent meta-analysis have been debated.,, It is important to remember, though, that the benefit from using metformin in the UKPDS was observed among patients with newly diagnosed type 2 diabetes, who would likely be at a lower absolute risk of diabetic complications than patients who have had the disease for a few years or have advanced renal or cardiovascular disease. Therefore, if anything, the absolute magnitude of the benefit of using metformin in “typical” patients with type 2 diabetes would likely be greater, because their initial absolute risk would be greater. The benefit of using metformin is fairly clear, but what is the risk of lactic acidosis among these patients? This is a more difficult question to answer. In a recent study by Salpeter and coauthors, pooled data from 194 studies showed no cases of lactic acidosis in over 35 000 patient-years of metformin use. It must be remembered that this condition occurs in diabetic patients independent of metformin use. In 1998 Brown and colleagues reported on the incidence of lactic acidosis before and after the introduction of the drug in the United States: they found no difference in the incidence rates in 41 000 patient-years. Before the drug's introduction, they placed the rate of lactic acidosis among patients not receiving metformin at 9.7–16.9 events per 100 000. The meta-analysis by Salpeter and coauthors suggests that the upper limit for the true incidence of lactic acidosis is 9.9 events per 100 000 patient-years among patients with type 2 diabetes not receiving metformin and 8.1 per 100 000 among those taking the drug. In a historical cohort, Stang and colleagues suggested a similar rate of lactic acidosis among metformin users at 9 cases per 100 000 patient-years. This suggests that diabetes alone is an equally relevant, if not more relevant, risk factor for lactic acidosis than is metformin use. In the trials included in the meta-analysis by Salpeter and coauthors, 16% of the patients were over the age of 65 years, and 44% of the trials allowed for the inclusion of patients with renal insufficiency, usually defined as a serum creatinine level of more than 132 mmol/L. Of the studies, 96% allowed for the inclusion of patients with hypoxemic comorbidities such as renal insufficiency, cardiovascular disease, liver disease or pulmonary disease; however, these authors were unable to determine how many patients with these conditions were included in the clinical trials, and so it is not possible to determine the specific incidence of lactic acidosis among patients with these supposed contraindications. Nonetheless, the incidence of lactic acidosis in these studies was still zero. Case studies have shown an association between reduced renal function, as indicated by elevated serum creatinine levels, and the incidence of lactic acidosis.,,, Sulkin and coauthors described 2 patients with reduced renal function (serum creatinine levels 150 and 203 mmol/L) in whom lactic acidosis developed while they were taking metformin. Misbin and colleagues found that, of 20 patients who died of lactic acidosis while taking metformin, 80% had significant renal impairment; the authors stated that lactate accumulation can result from renal impairment and its association with metformin use may have been coincidental. In addition, metformin-associated lactic acidosis can develop in patients with normal serum creatinine levels. There are published reports of 3 patients in whom lactic acidosis developed without there having been any indications of conditions that would increase the risk of this complication., All of the patients' hepatic, cardiac and renal functions were within normal limits, and all of the patients were taking normal doses of metformin (500 mg, 850 mg and 1000 mg, respectively, twice daily). In 1 case, the woman was underweight and died; possible intentional overdose of metformin could not be ruled out. In the other 2 cases, metformin therapy was stopped, and the patients recovered., An earlier case study described 6 patients with lactic acidosis whose serum creatinine levels ranged from 90 to 274 mmol/L before diagnosis and from 168 to 663 mmol/L after diagnosis; however, 3 of the patients, whose creatinine levels were normal, had undergone treatment involving contrast media. This may have been the reason for the development of lactic acidosis, although there is a lack of evidence to support a recommendation to withhold metformin therapy after the use of contrast media in patients with normal renal function. When the 79 cases of lactic acidosis in the literature are analyzed together, a unifying theme appears to emerge. Studies have shown that plasma metformin levels were not correlated to blood lactate levels, which raises the question of whether metformin is a causal factor in lactic acidosis., Lactic acidosis is associated with acute events such as myocardial infarction or congestive heart failure, acute renal insufficiency and sepsis. These 79 cases resulted after an estimated 1.1 million patient-years of metformin use, and in 37% of the cases the patient had a cardiac condition (myocardial infarction or congestive heart failure), in 24% an acute renal condition, in 8% an acute hepatic condition and in 4% sepsis. In all of these cases the underlying conditions could have themselves caused the lactic acidosis, so it is difficult to identify the degree to which metformin was responsible. There is little, if any, research exploring rates of lactic acidosis among metformin users with rates among users of other oral hypoglycemic agents. It appears that the incidence of metformin-associated lactic acidosis is not much different than the baseline incidence among people with type 2 diabetes. Even if the presence of renal failure increased the incidence of lactic acidosis 10-fold over baseline, the incidence would still be only about 1% over 10 years (with a baseline rate of 10 per 100 000 patient-years, the 10-fold increase would result in a rate of 100 per 100 000 patient-years, which would be about 1 per 1000 patient-years, or 1% over 10 years). The benefit of using metformin would be a reduction of 5% in diabetes-related deaths, 7% in all-cause mortality, 6% in myocardial infarction and 3% in stroke. Given the above information, it is clear that, even among patients with the so-called “contraindications,” the magnitude of the benefit of metformin therapy would clearly outweigh any potential risk. What, then, should clinicians do about this issue? They need to decide whether the above evidence warrants a discussion about the specific risk of lactic acidosis. If the issue of lactic acidosis is of enough concern to them, this issue should be discussed not only with patients taking metformin, but also with any patient who has type 2 diabetes, because the risk is really not a lot different. Perhaps a reasonable approach is to tell patients, as one should when prescribing any drug therapy, that if any unexplained symptoms develop such as nausea or vomiting, abdominal pain, rapid breathing, difficulty breathing, chest pain, weakening of the muscles in the legs and arms, diarrhea, skin rash and confusion, one of the potential reasons for these symptoms could be the drug but it could also possibly mean another medical condition. Regardless, if these symptoms occur, patients should seek prompt medical attention. Using metformin in a patient of advanced age (≥ 80 years) or in a patient who has reduced renal function requires one to consider the potential for decreased elimination ability. Unfortunately for metformin, there really is no solid evidence to guide clinicians as to what to do. A reasonable approach might be, as with all patients, to start with a low dose (250 mg twice daily) and increase the dose weekly, based on tolerance and effect of the drug on the surrogate end point of blood glucose level, to a maximum dose of 2000– 2500 mg/d. Based on general pharmacokinetic principles, rather than avoid the drug completely, it would be reasonable to reduce the maximum dose by about 50% in a patient with an estimated creatinine clearance of less than 1.0 mL/s. In conclusion, the evidence at present suggests that the use of metformin in patients who are over the age of 80 years, have congestive heart failure or have renal insufficiency leads to a benefit that far outweighs the potential harm. We would suggest that it is a “contraindication” to not use metformin in people with type 2 diabetes with these contraindications. β See related article page 505 FootnotesIn this first of 2 commentaries, James McCormack and coauthors present their view on whether the current contraindications to metformin therapy are warranted for certain patients with type 2 diabetes. We sent the article, which arrived unsolicited, out for peer review and received a particularly thoughtful assessment from George Fantus, who weighed some of the same evidence and reached a different conclusion (see page 505). We thought you might find it as interesting as we did to read 2 careful considerations of the same clinical question that reach different answers. Evidence, at least for this issue, is in the eye of the beholder. — CMAJ This article has been peer reviewed. Contributors: Hugh Tildesley, the chief investigator of the project, drafted portions of the article, provided relevant background information, contributed intellectual content and gave final approval. Kevin Johns performed the literature search, was the primary author of the article and gave final approval. James McCormack, the co-investigator of the project, interpreted data, drafted portions of the article, critically revised the article for important intellectual content and gave final approval. Competing interests: None declared. Correspondence to: Dr. James McCormack, Endocrine Research, St. Paul's Hospital, 467–1081 Burrard St., Vancouver BC V6Z 1Y6; fax 604 631-5154;ac.cbu.egnahcretni@amroccmj References
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Articles from CMAJ : Canadian Medical Association Journal are provided here courtesy of Canadian Medical Association
Published online 2011 May 20. doi: 10.2337/dc10-2361
PMID: 21617112
This article has been cited by other articles in PMC.
A common clinical conundrum faces all U.S. practitioners treating patients with type 2 diabetes. Today’s U.S. Food and Drug Administration prescribing guidelines for metformin contraindicate its use in men and women with serum creatinine concentrations ≥1.5 and ≥1.4 mg/dL (≥132 and ≥123 µmol/L), respectively. In a patient tolerating and controlled with this medication, should it automatically be discontinued as the creatinine rises beyond these cut points over time? Stopping metformin often results in poorly controlled glycemia and/or the need for other agents with their own adverse-effect profiles. Moreover, is the now widespread use of estimated glomerular filtration rate (eGFR) in lieu of serum creatinine levels creating even more confusion, especially in those with abnormalities in one but not the other indirect measure of renal function? Indeed, more than a decade and a half after metformin became available in the U.S., debate continues over the best approach in these settings (–). How many patients are unable to receive this medication on the basis of guidelines which, although well intentioned, are somewhat arbitrary and outdated based on modern assessments of renal status? ADVANTAGES OF METFORMINThere is some evidence that early treatment with metformin is associated with reduced cardiovascular morbidity and total mortality in newly diagnosed type 2 diabetic patients (). However, the data come from a small subgroup of a much larger trial. In contrast, despite multiple trials of intensive glucose control using a variety of glucose-lowering strategies, there is a paucity of data to support specific advantages with other agents on cardiovascular outcomes (–). In the original UK Prospective Diabetes Study (UKPDS), 342 overweight patients with newly diagnosed diabetes were randomly assigned to metformin therapy (). After a median period of 10 years, this subgroup experienced a 39% (P = 0.010) risk reduction for myocardial infarction and a 36% reduction for total mortality (P = 0.011) compared with conventional diet treatment. Similar benefits were not observed in those randomly assigned to sulfonylurea or insulin. In an 8.5-year posttrial monitoring study, after participants no longer were randomly assigned to their treatments, individuals originally assigned to metformin (n = 279) continued to demonstrate a reduced risk for both myocardial infarction (relative risk 33%, P = 0.005) and total mortality (relative risk 27%, P = 0.002) (). The latter results are even more impressive when one considers that HbA1c levels in all initially randomly assigned groups had converged within 1 year of follow-up. Unlike sulfonylureas, thiazolidinediones, and insulin, metformin is weight neutral (), which makes it an attractive choice for obese patients. Furthermore, the management of type 2 diabetes can be complicated by hypoglycemia, which can seriously limit the pursuit of glycemic control. Here, too, metformin has advantages over insulin and some types of insulin secretagogues; by decreasing excess hepatic gluconeogenesis without raising insulin levels, it rarely leads to significant hypoglycemia when used as a monotherapy (,). As a result, metformin is widely considered an ideal first-line agent for the treatment of type 2 diabetes, as recommended by several clinical guidelines (–14). In addition to such benefits, metformin reduces the risk of developing diabetes in individuals at high risk for the disease () and has been considered as a reasonable “off-label” approach in selected individuals for diabetes prevention (). HISTORICAL PERSPECTIVEDespite these proven benefits, metformin remains contraindicated in a large segment of the type 2 diabetic population, largely because of concerns over the rare adverse effect of lactic acidosis. For these reasons, the drug has been restricted to individuals with normal creatinine levels as a surrogate for renal competence. Other contraindications (e.g., any significant hypoxemia, alcoholism, cirrhosis, a recent radiocontrast study) also increase the risk for or the consequences of lactic acidosis, but these are not the topic of this review. Metformin belongs to the biguanide drug class (previous members include phenformin and buformin), developed for lowering glucose in the 1950s. Initial enthusiasm for biguanides was tempered over the next two decades by the growing recognition of their risk of lactic acidosis. A marked reduction in biguanide use occurred in the mid-1970s because phenformin, extensively adopted in clinical practice, was implicated in a number of fatal cases of this severe metabolic decompensation (). The association with lactic acidosis eventually led to its withdrawal from the market. Importantly, lactic acidosis with phenformin seems to occur ~10–20 times more frequently than with metformin (). In contrast to metformin, modestly raised phenformin concentrations may reduce peripheral glucose oxidation and enhance peripheral lactate production, which can increase circulating lactate levels. In fact, phenformin levels correlate with lactate concentration, whereas metformin levels do not (). In addition, ~10% of European Caucasians have an inherent defect in phenformin hydroxylation, which may lead to drug accumulation and, as a result, elevated lactate levels (). The experience with phenformin resulted in cautious use of metformin in Europe. In the 1980s, the creatinine cut points for contraindication to metformin were considered to be appropriate at 1.4 mg/dL in women and 1.5 mg/dL in men. This was based on the calculated ability to remove 3 g of metformin (an amount slightly beyond the maximum daily U.S. dose) at steady-state levels within 24–48 h. In fact, the ability to comfortably remove the drug extends up to creatinine levels of 1.8–2.0 mg/dL, but the cut points chosen were intentionally set lower to ensure that those patients who may be lost to follow-up and whose creatinine levels increase over time would not be at risk for appreciable drug accumulation. Metformin was not approved in the U.S. until December of 1994 and was marketed in 1995. The experience with phenformin led to judicious labeling and recommendations for careful monitoring of adverse events. Its new-drug application was the subject of extensive review. The risk for lactic acidosis was estimated to be ~3 cases per 100,000 patient-years based on cases reported from France, the U.K., and other countries where pharmacovigilance information was available (). Similar estimates were quoted from Sweden (2.4 cases per 100,000 patient-years), where the number of cases appeared to be decreasing despite rising clinical use of metformin. After careful deliberation, metformin was approved by the U.S. Food and Drug Administration, with lactic acidosis risk seen as small, although inevitable with future widespread availability of the drug. CLINICAL PRACTICE GUIDELINESThe prescribing information for metformin in the current label is explicit with respect to renal contraindications, based on serum creatinine cut points. It proscribes use at or above the 1.4 and 1.5 mg/dL levels in women and men, respectively. The recently updated Kidney Disease Outcomes Quality Initiative guidelines from the National Kidney Foundation are perfectly consistent with the label (22). Yet, certain U.S. practice guidelines substantially differ in their recommendations for metformin use related to renal status. The annually updated clinical practice guidelines issued by the American Diabetes Association, for example, do not actually discuss renal thresholds () but refer the reader to a separate consensus statement for prescribing details. This statement, authored by members of the American Diabetes Association and European Association for the Study of Diabetes, reports that metformin seems safe unless eGFR falls to <30 mL/min per 1.73 m2 (). Clinical guidelines outside of the U.S. incorporate the eGFR for determination of metformin safety. In the U.K., for example, prescribing guidelines consider both creatinine and eGFR for assessing treatment eligibility. The National Institute for Health and Clinical Excellence recommends reviewing the clinical circumstances when serum creatinine exceeds 130 µmol/L (1.5 mg/dL) or eGFR falls below 45 mL/min per 1.73 m2. The National Institute for Health and Clinical Excellence further specifies that metformin be stopped if serum creatinine exceeds 150 µmol/L (1.7 mg/dL) (a higher threshold than in the U.S.) or eGFR is below 30 mL/min per 1.73 m2 (14). In contrast, the Canadian Diabetes Association practice guidelines are now based solely on eGFR, recommending caution with eGFR <60 mL/min per 1.73 m2 and contraindicating its use with eGFR <30 mL/min per 1.73 m2 (23). The Australian Diabetes Society practice guidelines similarly recommend against metformin with eGFR <30 mL/min per 1.73 m2 and caution with eGFR 30–45 mL/min per 1.73 m2 (24). Thus, although there is clear recognition that renal failure may be a risk factor for adverse events with metformin use, there is significant divergence in opinion across the globe regarding the optimal definition of safety. METFORMIN PHARMACOKINETICSThe principal reason for carefully setting renal thresholds is that metformin is eliminated unchanged primarily by the kidneys. Thus, one of the most important risk factors for elevated metformin concentrations (which are proposed to lead to lactic acidosis) is the inability to clear the drug efficiently. Metformin has a 50–60% bioavailability and is absorbed mainly in the small intestine. It does not appear to bind appreciably to plasma proteins. The maximum plasma concentration is observed ~2 h after oral dosing, typically reaching a Cmax of 1–2 µg/mL (~10 µmol/L). Metformin accumulates in the walls of the small intestine and salivary glands as well as in the kidney (). It has a plasma elimination half-life of 6.2 h and is renally eliminated both by filtration and active tubular secretion (). In careful experiments, Tucker et al. () studied metformin kinetics in 4 healthy subjects and 12 type 2 diabetic subjects and found that plasma renal clearance of metformin highly correlated with creatinine clearance (CrCl; r = 0.85, P < 0.001). However, the relationship between physiological clearance of an actual oral dose and CrCl was much weaker (r = 0.66, P < 0.01). Therefore, the investigators postulated that other factors may impact this relationship, such as perhaps gastrointestinal absorption of metformin in patients with renal failure and/or nonrenal clearance of a small amount of the drug. In another pharmacokinetic study (), a single 850-mg dose of metformin was given to 21 healthy subjects and 13 subjects with renal insufficiency (mild to severe). In the control group (data presented are mean ± SD) (mean CrCl 112 ± 8 mL/min), average renal metformin clearance was 636 ± 84 mL/min, whereas in mild chronic kidney disease (CKD) (CrCl 61–90 mL/min; mean 73 ± 7) clearance was reduced at 384 ± 122 mL/min. The mean renal clearance of metformin was lower in subjects with moderate (CrCl 31–60 mL/min; mean 41 ± 9) and severe (CrCl 10–30 mL/min; mean 22 ± 6) CKD, measuring 108 ± 57 and 130 ± 90 mL/min, respectively. Similarly, maximum concentration and the area under the concentration time curve were increased in individuals with moderate to severe CKD compared with those with mild CKD or normal renal function. Based on the regression analysis, both CrCl and age were found to be important predictors of metformin clearance. This study did not provide evidence for specific thresholds at which lactate production may begin to rise. These reports have relied on information derived from single doses of metformin, which may not reflect chronic-treatment pharmacokinetics. In contrast, few reports have assessed the impact of renal insufficiency on metformin clearance during long-term use. Indeed, one such study () concluded that metformin can be efficiently cleared in mild-to-moderate CKD. In this investigation, 24 older patients (aged 70–88 years) were administered metformin 850 mg/day or 1,700 mg/day based on CrCl of 30–60 mL/min (n = 11) or >60 mL/min (n = 13), respectively. After 2 months, metformin remained in the therapeutic range and lactate within the reference limits in all participants. In addition, the measured levels of metformin and lactate were not statistically different between those with and without renal impairment (). Another recent study () evaluated metformin levels in patients with type 2 diabetes and varying renal function. GFR was estimated based on cystatin C levels. The median dose of metformin was 1,500 mg/day. The median serum level of metformin was 4.5 µmol/L (~0.6 µg/mL) (range 0.1–20.7) in patients with eGFR >60 mL/min per 1.73 m2 (n = 107), 7.7 µmol/L (~1.0 µg/mL) (0.1–15.2) with eGFR 30–60 mL/min per 1.73 m2 (n = 21), and 8.9 µmol/L (~1.1 µg/mL) (6.0–18.6) with eGFR <30 mL/min per 1.73 m2 (n = 9). Notably, there were wide variations in these levels within each group, with few patients having serum levels >20 µmol/L (>~2.6 µg/mL). However, the “unsafe” metformin concentration is not really known. At usual clinical doses and schedules, steady-state plasma concentrations are generally <1 µg/mL (<7.8 µmol/L). Maximum plasma levels during controlled clinical trials do not generally exceed 5 µg/mL (38.8 µmol/L), but these have not typically enrolled CKD patients. Moreover, whether measurement of metformin levels actually can aid in the prediction of lactic acidosis risk remains unclear. Therefore, although these studies provide some information on the relationship between renal function and metformin concentrations, they do not clarify the issue of toxicity and lactic acidosis risk. Many of the early pharmacokinetic studies with metformin actually relied on CrCl based on 24-h urine collection for creatinine. How well the current serum creatinine cut points reflect the ability to effectively clear the drug also is unknown. Creatinine levels, in general, vary inversely with GFR. However, important limitations to the estimation of renal function with creatinine should be considered. First, serum creatinine can only be used reliably in patients with stable kidney function. Second, variation in creatinine production may differ among and within individuals over time, especially if there are significant changes in muscle mass or in physical activity. Variability in creatinine secretion, extrarenal creatinine excretion, assay method, and equipment can all affect serum measurements. Calculated estimates (clearance from the Cockroft-Gault and eGFR from the Modification of Diet in Renal Disease equation) have been developed to incorporate known demographic and clinical factors affecting serum concentrations. These equations have their own inherent shortcomings, such as residual limitations with respect to age and race, underestimation of GFR in the context of diabetic renal disease (Cockroft-Gault and Modification of Diet in Renal Disease) (), and in obese individuals (Modification of Diet in Renal Disease) (). However, they provide better estimation of renal function than creatinine alone. Moreover, development of new estimating equations, such as the Chronic Kidney Disease Epidemiology Collaboration equation, may allow for even more accurate estimates of renal function in the future. Finally, dosing considerations by the Food and Drug Administration for other medications (e.g., sitagliptin, fenofibrate) are generally based on CrCl estimated from such calculations and not on creatinine levels themselves. LACTIC ACIDOSIS ASSOCIATED WITH METFORMIN THERAPYEven though elevated metformin concentrations have been proposed to lead to lactic acidosis, there are few data regarding the level predisposing to hyperlactatemia. In fact, multiple studies suggest that elevated circulating lactate levels, often attributed to metformin, may actually not be caused by the drug. First, lactic acidosis occurs in patients with type 2 diabetes more frequently than in the general population; in some reports, the observed rate appears to be similar in patients on metformin versus other glucose-lowering agents (). Second, metformin and lactate levels do not necessarily appear to correlate, such that higher metformin concentrations do not consistently occur in those with more severe degrees of lactic acidosis (,). Finally, metformin levels are not linked to mortality in those who develop lactic acidosis, perhaps reflecting the primary effect of the underlying cause of the acidosis (e.g., hypoxia, hemodynamic compromise) on outcomes rather than incriminating metformin itself (–). Although lactic acidosis remains a recognized, albeit rare, adverse event associated with metformin, the number of lactic acidosis cases continues to be very small, particularly when one considers the widespread use of this drug. In the largest updated Cochrane meta-analysis, Salpeter et al. () pooled data from 347 comparative trials and cohort studies. Not a single case of lactic acidosis was found in >70,000 metformin patient-years or >55,000 nonmetformin person-years. In this analysis, 53% of prospective studies allowed for inclusion of renal insufficiency, but patient-level serum creatinine concentrations were not available for review. Based on statistical inference, the estimated upper limit of true incidence was 4.3 and 5.4 cases per 100,000 patient-years in the metformin and nonmetformin groups, respectively. This investigation suggests that lactic acidosis is extremely rare and the incidence does not differ in those treated with metformin versus other agents. In a large, nested, case-control analysis of the U.K. general practice research database (), the crude incidence of lactic acidosis was even lower at 3.3 cases per 100,000 person-years among metformin users and 4.8 cases per 100,000 person-years among sulfonylurea users (in very close agreement to the estimates of 3 and 2.4 cases per 100,000 patient-years from Europe and Scandinavia before metformin’s U.S. approval). Given all of these findings, some () have argued that the occurrence of lactic acidosis with metformin use is merely coincidental and that there is no tangible evidence from prospective observational studies or clinical trials that the drug increases its incidence. Of course, all these data have been collected in the context of contemporaneous strict metformin-prescribing guidelines. It is possible that looser restrictions may have led to more frequent occurrence of lactic acidosis. In summary, lactic acidosis remains exceedingly rare in clinical trials and cohort studies of metformin therapy. Moreover, the available data suggest that lactate levels and risk of lactic acidosis do not differ appreciably in patients taking this versus other glucose-lowering agents. Thus, the long-proclaimed causal relationship between metformin and lactic acidosis remains in question. CURRENT USE OF METFORMIN IN CKDGiven the current contraindications in the U.S., some might consider it a challenge to conduct a new clinical trial to evaluate the use of metformin in individuals with various degrees of impaired renal function, taking into account new criteria for assessing glomerular filtration. Yet, evidence suggests that metformin is often already used in practice outside of the current labeling contraindications, prescribed in full knowledge of the relevant cutoffs (–). For example, in a review () of restrictions to metformin therapy conducted in Scotland, 24.5% of metformin users had filled a prescription despite active contraindications (3.4% had the specific local exclusion of a serum creatinine ≥1.7 mg/dL recorded twice on different days within 4 weeks). A single case of lactic acidosis during 4,600 patient-years of follow-up occurred in a patient with an extensive acute myocardial infarction who developed acute renal failure and died the same day. Given the clinical scenario, the authors intimated that acidosis had occurred because of hemodynamic compromise related to the infarct and not to metformin accumulation. In a U.S. study () performed in the primary care practice setting, 4.5% of patients treated with metformin had creatinine levels >1.4 and 1.5 mg/dL in women and men, respectively. Two other studies (,) of sicker patients admitted to hospitals in Germany and the U.S. confirmed high frequency of metformin use despite various contraindications (73 and 27%, respectively). When one considers the imperfect reflection of actual renal function by serum creatinine, metformin is likely used even more frequently in patients with impaired GFR than that suggested by the above studies. In the aforementioned U.S. primary practice setting, where 4.5% of patients were given metformin despite creatinine-based contraindications, 17.7% of women and 13.4% of men receiving metformin actually had an abnormally low eGFR (≤60 mL/min per 1.73 m2) (). Likewise, in another single U.S. center cross-sectional study (), 15.3% of patients with type 2 diabetes and eGFR <60 mL/min per 1.73 m2 were receiving metformin. Such frequent “inappropriate” use of metformin in patients is further suggested by data from the National Health and Nutrition Examination Survey (1999–2006) (). Among individuals with eGFR <60 mL/min per 1.73 m2 and diabetes, 32.2% were treated with metformin and had a normal creatinine level (<1.5 mg/dL), whereas 13.4% were treated with metformin despite a frankly elevated creatinine level (>1.5 mg/dL). The use of metformin in mild-to-moderate CKD clearly is not at all uncommon. Two studies have attempted to translate creatinine into corresponding eGFR cut points in the context of metformin therapy. In a review () of prescribing practices in the U.K., appropriate use of the drug was defined on the basis of creatinine level ≤1.7 mg/dL. Of 11,297 patients meeting those criteria, 82% had an eGFR <90, 25.5% <60, and 2.8% <30 mL/min per 1.73 m2. The authors calculated that the eGFR threshold of 36 mL/min would result in a similar number of patients becoming ineligible for metformin compared with the serum creatinine threshold of 1.7 mg/dL (although some patients would become newly eligible and some who previously qualified would now become ineligible). The authors proposed that if the current practice is considered safe (and based on the review by Salpeter et al. [], this appears to be so), then a switch to an eGFR-based cut point may be both a more practical and a more accurate way to limit metformin access in those with significantly impaired renal function. In another British study of 12,482 patients with diabetes, an eGFR cutoff of 41 mL/min per 1.73 m2 in men and 30 mL/min per 1.73 m2 in women resulted in a similar proportion of patients having metformin withheld compared with the serum creatinine threshold of 1.7 mg/dL (). The investigators therefore proposed the pragmatic eGFR limit of 30 mL/min per 1.73 m2 to denote absolute contraindication to therapy. Limited data specifically address metformin’s long-term safety in patients with mild-to-moderate renal failure (–). These studies found no increased risks in various degrees of renal insufficiency but were limited by small size and significant methodological shortcomings. Recently, an analysis of the Reduction of Atherothrombosis for Continued Health Registry suggests that the proposed cardiovascular benefits of metformin may extend to patients with established atherosclerosis and moderate CKD (). In this large, observational study of >19,000 subjects with a history of atherothrombotic disease, 1,572 patients were using metformin with eGFR 30–60 mL/min per 1.73 m2. After adjustment for baseline factors and propensity score, metformin use was associated with a significant reduction in 2-year mortality in the overall population (hazard ratio 0.76 [95% CI 0.65–0.89]), including in those with moderate CKD (0.64 [0.48–0.86]). However, lack of information with respect to the duration of metformin use and HbA1c, as well as the observational nature of the study, require further confirmation of the mortality benefit in similar patient cohorts in prospective trials. Although these data are reassuring, we must note that there are no randomized clinical trials that specifically evaluated the safety of metformin use and potential cardiovascular benefits in patients with CKD. PROGRESSION OF CKD IN PATIENTS WITH DIABETESRenal function is dynamic, and renal dysfunction in diabetes is typically progressive (50). Even in the absence of an acute event, glomerular function slowly declines with aging as nephron mass is lost. The renal thresholds for the acceptability of metformin therapy should therefore ideally account for the tempo of CKD progression. The assessment of renal function in clinical practice occurs periodically, and the degree of renal dysfunction may change appreciably between these assessments. Therefore, it is important to know how quickly GFR declines in the typical spectrum of nephropathy among diabetic patients, particularly when considering metformin therapy. Few studies have, however, systematically evaluated the rate of progression of renal dysfunction in the general diabetic patient population by directly measuring GFR over time. Some data suggest that eGFR tends to underestimate the rate of GFR decline by as much as 28% when compared with direct measurement (). Nevertheless, most of the available data are based on estimations. A recent British population-based cohort study () of 3,431 diabetic patients examined renal decline as measured by changes in eGFR (Modification of Diet in Renal Disease). The analysis of data collected over 7 years demonstrated that the rate of progression was lowest among individuals who were normoalbuminuric (0.3% or ~0.2 mL/min per 1.73 m2 decline in eGFR per year), intermediate in those with microalbuminuria (1.5% or ~1.2 mL/min per 1.73 m2 per year), and highest in those with macroalbuminuria (5.7% or 4.5 mL/min per 1.73 m2 per year). In a large Dutch study (), eGFR fell by 0.5 mL/min per 1.73 m2 per year in the general population but to a greater extent in those with hypertension, diabetes, and macroalbuminuria (1.9 mL/min per 1.73 m2 per year). Based on these and other results (), average annual progression of renal dysfunction in diabetes appears to be in the range of −1 to −4 mL/min per 1.73 m2 of filtration capacity, dependent in part on other risk factors and the use of renoprotective therapies. The decline is slow, but, importantly, the majority of patients in these studies had normal kidney function at the outset. There is less information available on diabetic patients with CKD. It should also be noted that diabetes places an individual at increased risk for other causes of renal disease (). Thus, all diabetic patients, especially those with CKD, may be at risk for more rapid decline in their renal function or acute kidney injury. Despite these appropriate concerns, most of the available data would suggest that, on average, eGFR declines slowly in diabetes, although it can be accelerated to some degree in the presence of albuminuria. If eGFR is calculated annually (and more frequently in those at high risk for deterioration in renal function), it is unlikely that patients will experience changes in their eGFR levels large enough to rapidly alter the safety of metformin therapy. CONCLUSIONS AND RECOMMENDATIONSAlthough metformin is eliminated renally, and accumulation may conceivably lead to lactic acidosis, there currently is limited systematic evidence to substantiate continued reliance on the creatinine-based contraindications in use in the U.S. Indeed, in the modern era of eGFR, this measure of glomerular filtration appears to give a more reliable estimate of renal dysfunction. Metformin-associated lactic acidosis is exceedingly rare based on the available literature, and even though the use of metformin has not been comprehensively assessed in individuals with CKD, there is extensive evidence that this agent often is used without adverse effects in those with mildly to moderately reduced renal function. In the context of rising concerns regarding other glucose-lowering therapies (55), safety restrictions over the use of metformin in this population may result in the drug being stopped prematurely and unnecessarily in some patients. This may lead to significant deterioration in glycemic control and/or the need for other glucose-lowering medications, which present their own safety concerns. An evidence-based approach to prescribing metformin in this group appears warranted, taking into account the current pervasive use of eGFR in clinical care. We therefore suggest that the current guidelines for metformin use in the U.S. be updated. These recommendations should include eGFR thresholds that are generally consistent with the National Institute for Health and Clinical Excellence guidelines in the U.K. and those endorsed by the Canadian Diabetes Association and the Australian Diabetes Society (Table 1). Metformin may be continued (or initiated) with eGFR <60 mL/min per 1.73 m2, but renal function should be monitored closely (every 3–6 months). The dose of metformin should be reviewed and reduced (e.g., by 50% or to half-maximal dose) in those with eGFR <45 mL/min per 1.73 m2, and renal function should be monitored closely (every 3 months). Metformin should not be initiated in patients at this stage, however. The drug should be stopped once eGFR falls to <30 mL/min per 1.73 m2. Additional caution is required in patients with anticipated significant fluctuations in renal status or those at risk for abrupt deterioration in kidney function, based on previous history, other comorbidities, albuminuria, and medication regimen (e.g., potent diuretics or nephrotoxic agents). Table 1Proposed recommendations for use of metformin based on eGFR
Additional caution is required in patients at risk for acute kidney injury or with anticipated significant fluctuations in renal status, based on previous history, other comorbidities, or potentially interacting medications. Without question, such a treatment program could not be implemented without meticulous clinical follow-up, clear communication with patients regarding risks and benefits of therapy, and adherence to frequent monitoring. The plan, therefore, should be modified in patients with suboptimal adherence to medical instructions or regular follow-up. It is clear that vigilance would be required so that cases of lactic acidosis do not emerge because of inappropriate use of metformin in patients with more advanced and/or unstable CKD. Given the frequency with which clinicians must decide whether to continue or initiate metformin in mild-to-moderate CKD, these recommendations would have a profound impact on clinical practice. Such proposals, being consistent with the sentiments of several well-respected national guideline committees, should be reviewed by professional medical organizations in the U.S., such as the American Diabetes Association. If a consensus emerges, perhaps the Food and Drug Administration might reconsider the current metformin-prescribing guidelines, which, like those of other nonbranded compounds, tend to remain static despite emerging evidence and changes in clinical care. In the future, more research in this important area is needed, including prospective, randomized trials of metformin at varying degrees of renal impairment and/or closer examination of registries of CKD patients receiving metformin. AcknowledgmentsNo potential conflicts of interest relevant to this article were reported. K.J.L. researched data, wrote the manuscript, reviewed and edited the manuscript, and contributed to discussion. C.J.B. and S.E.I. reviewed and edited the manuscript and contributed to discussion. References
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Articles from Diabetes Care are provided here courtesy of American Diabetes Association
The GPS/GNSS/Iridium antenna is a folded Quadrifilar Helix Antenna (QHA) with novel features to improve frequency coverage compared to existing QHA designs. The GNSS frequencies covered include modernized GPS (L1, L2, L5), GLONASS, Galileo, and Beidou (Compass), spanning from 1164 to 1300 MHz and 1559 to 1611 MHz. Self-phased Quadrifilar Helix on a Solid Core A selection of helix antennas in Antenna Magus. Up until release 2.3, Antenna Magus has provided information, models and designs for 8 different helical antennas in the basic helix antenna classes, namely: axial-mode helices, normal-mode helices and quadrifilar helices.
Design of Quadrifilar Helix Antenna with Parasitic Element and Channel Characterisation for Small Cell Network Tengku Faiz Bin Tengku Mohmed Noor Izam
Submitted for the Degree of Doctor of Philosophy from the University of Surrey Institute for Communication Systems Department of Electronic Engineering Faculty of Engineering and Physical Sciences University of Surrey Guildford, Surrey GU2 7XH, UK August 2015 ©Tengku Faiz 2015 Abstract Small cell networks (SCN) have emerged as a viable solution for improving the spectral efficiency in order to satisfy the growing demand for high data rate mobile network. SCNs consist of multiple short range base station (BS) to cover small areas. The BS is typically known as femtocell BS. Polarisation mismatch loss between the BS and mobile station (MS), and inter-cell interference between BSs can be the performance limiting factors for SCN deployment in non-cluttered open space. This work covers the antenna design for the femtocell BS and channel characterisations within a SCN environment. Two designs for quadrifilar helix antenna (QHA) gain improvement using parasitic loop have been proposed. The designs are based on parasitic meandered loop (PML) and parasitic quadrifilar helix loop (PQHL). These parasitic loops are able to improve the boresight gain by up to 1.8 dB. Another design that is evaluated in this work is the switched parasitic QHA (SPQHA). By using parasitic elements at the side of the QHA, it gives a low complexity beam steering capability with up to 35° beam tilt. This feature is useful in cooperative SCNs to improve coverage and minimise interference. The performance of BS antennas with different polarisations against mobile station (MS) under random human handling in a real environment has been evaluated. Results show that polarisation mismatch between the BS and MS can be severe due to lack of signal depolarisation in short range communication. Results also show that a circular polarised BS antenna can be a good compromise to minimise polarisation mismatch loss in a SCN environment. A second field measurement has been conducted to evaluate the performance of the SPQHA in a real environment. Results have shown that SPQHAs are able to provide a high diversity gain. With local parasitic switching on one BS, 8 dB diversity gain can be achieved. With global parasitic switching on two BSs, 13 dB diversity gain is obtained. Furthermore, MIMO antenna selection using SPQHAs has also been shown to be able to match the performance of a 8-elements QHA-based MIMO setup. As a result, MIMO SPQHA can reduce the number of RF-chains required as compared to a full MIMO setup. Acknowledgements I would to thank my supervisor, Dr Timothy Brown who has been patiently guiding and advising me in completing this research. His expertise and guidance in the research area has been a tremendously help. I also would like to express my gratitude to the Government of Malaysia and University Malaya which have provided me the opportunity to embark in this research by providing a full sponsorship. Finally, praised to Almighty God who has blessed me with the courage and endurance in completing this interesting research. Contents Contents iii List of Figures vi List of Tables x Nomenclature xiv 1 2 Introduction 1.1 Motivation . . . . . . . 1.2 Contributions . . . . . 1.3 Publication . . . . . . 1.4 Structure of the Thesis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Literature Review 2.1 Small Cell Network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.1 Femtocell . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.2 Public Femtocell . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.3 Interference in Two-tier Network . . . . . . . . . . . . . . . . . . 2.1.4 Antenna Techniques for Capacity Improvement and Interference Mitigation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2 Operator’s Influence On Mobile Phone Antenna . . . . . . . . . . . . . . . 2.3 Signal Polarisation and Cross-polarisation discrimination (XPD) . . . . . . 2.3.1 Signal Polarisation in Free Space . . . . . . . . . . . . . . . . . . 2.3.2 Cross-Polarisation Discrimination (XPD) . . . . . . . . . . . . . . 2.4 Base Station Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4.1 Antenna Elements . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4.2 Reconfigurable Antenna . . . . . . . . . . . . . . . . . . . . . . . 1 1 2 3 3 5 5 5 6 8 9 10 12 12 14 16 16 18 iv Contents 2.5 2.6 3 4 5 2.4.3 Parastic Antenna . . . . . . . . . Quadrifilar Helix Antenna (QHA) . . . . 2.5.1 Introduction . . . . . . . . . . . . 2.5.2 Miniaturisation Techniques . . . . 2.5.3 Bandwidth and Gain Improvement 2.5.4 Multiband Capability . . . . . . . 2.5.5 Intelligent QHA (IQHA) . . . . . 2.5.6 Beamsteering Capability . . . . . Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Evaluation of Base Station Antenna with Different Polarisations in a Small Cell Network 3.1 Mobile stations’ Antennas . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2 The Effect of Human Body on the Mobile Stations’ Antennas Performance 3.3 Performance Measurement of BS Transmit Antenna with Different Polarisations in Anechoic Chamber. . . . . . . . . . . . . . . . . . . . . . . . . 3.3.1 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.4 Field Measurement Campaign . . . . . . . . . . . . . . . . . . . . . . . . 3.4.1 Measurement site . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.4.2 Measurement Setup . . . . . . . . . . . . . . . . . . . . . . . . . . 3.4.3 Result: Overall Characteristic . . . . . . . . . . . . . . . . . . . . 3.4.4 Result: Small Scale Fading (Fast Fading) . . . . . . . . . . . . . . 3.4.5 Large Scale (Slow Fading) Characteristic . . . . . . . . . . . . . . 3.5 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Controlling the Radiation Pattern of Quadrifilar Helix Antenna 4.1 Reference Quadrifilar Helix Antenna . . . . . . . . . . . . . . 4.2 Parasitic Loop . . . . . . . . . . . . . . . . . . . . . . . . . . 4.2.1 Parasitic Meandered Loop (PML) . . . . . . . . . . . 4.2.2 Parasitic Quadrifilar helix Loop (PQHL) . . . . . . . . 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) . . . . . . 4.4 Independent Feed QHA with Optimal Combining . . . . . . . 4.5 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 20 20 22 23 24 25 25 26 29 31 33 35 40 40 40 41 45 50 53 54 . . . . . . . 56 57 65 66 75 84 92 95 Side Parasitic and MIMO QHA Measurement Setup 5.1 Measurement Site . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97 97 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . v Contents 5.2 5.3 5.4 5.5 5.6 5.7 5.8 5.9 5.10 5.11 5.12 5.13 6 Antenna Configuration . . . . . . . . . . . . . . . . . . . . . . . 5.2.1 Transmit Antennas . . . . . . . . . . . . . . . . . . . . . 5.2.2 Receive Antennas . . . . . . . . . . . . . . . . . . . . . . 5.2.3 Cable Configurations . . . . . . . . . . . . . . . . . . . . Measurement Equipment . . . . . . . . . . . . . . . . . . . . . . 5.3.1 Elektrobit Propsound Wideband MIMO Channel Sounder 5.3.2 Equipment Parameter . . . . . . . . . . . . . . . . . . . . WLAN Interference . . . . . . . . . . . . . . . . . . . . . . . . . Measurement Scenarios . . . . . . . . . . . . . . . . . . . . . . . Post Processing . . . . . . . . . . . . . . . . . . . . . . . . . . . Path Loss Compensation From The Measured Data . . . . . . . . Antenna Selection Combining . . . . . . . . . . . . . . . . . . . Result I: Diversity Gain . . . . . . . . . . . . . . . . . . . . . . . 5.9.1 Local mean diversity . . . . . . . . . . . . . . . . . . . . 5.9.2 Instantaneous power diversity . . . . . . . . . . . . . . . Result II: Evaluation of the Downlink Interference . . . . . . . . . Result III: Comparison with QHA-based MIMO . . . . . . . . . . Result IV: Channel Characteristic . . . . . . . . . . . . . . . . . Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Conclusion and Further Work 6.1 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.2 Further work . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.2.1 Further design improvement for the Side Parasitic QHA 6.2.2 Nested Parasitic Quadrifilar Helix Loop (PQHL) . . . . 6.2.3 QHA Feed Network Improvement . . . . . . . . . . . . 6.2.4 Further Field Measurement for Markov Chain Parameter 6.2.5 Evaluation for Large Scale of SPQHA . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98 98 102 105 105 105 107 109 110 113 114 114 115 115 116 120 122 126 129 . . . . . . . 131 131 133 133 133 133 134 134 135 Appendix A Derivation of Polarisation Efficiency for transmission using LP and CP antennas 148 List of Figures 2.1 2.2 2.3 2.4 2.5 2.6 London Waterloo Station during peak hour (Example scenario). Mobile station orientations. . . . . . . . . . . . . . . . . . . . . Separation angle in linearly polarised transmission systems. . . . Polarisation efficiency vs. tilt angle for linear polarised antennas. Signal depolarisation in a real environment. . . . . . . . . . . . Printed quadrifilar helix antenna (QHA). . . . . . . . . . . . . . 3.1 3.2 3.3 3.4 3.5 3.6 Mobile station orientations (Reproduce in this section for reference). . . . . Mobile stations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Radiation pattern of the MS1 . . . . . . . . . . . . . . . . . . . . . . . . . . Radiation pattern of the MS2 . . . . . . . . . . . . . . . . . . . . . . . . . . Mobile stations’ return loss, S11 in various environments. . . . . . . . . . . MS1 radiation pattern without and with human interaction on the φ plane at θ = 90° in Talk (α = 60°) and Data (β = 25°) Modes. . . . . . . . . . . . MS2 radiation pattern without and with human interaction on the φ plane at θ = 90° in Talk (α = 60°) and Data (β = 25°) Modes. . . . . . . . . . . . Measured MS antennas gain in the respective modes and directions without (in free space) and with human interaction. . . . . . . . . . . . . . . . . . . Measurement site at the School of Management building, University of Surrey. Square patch antenna photo and radiation patterns. . . . . . . . . . . . . . QHA photograph and radiation pattern. . . . . . . . . . . . . . . . . . . . Propsound MIMO Channel Sounder setup. . . . . . . . . . . . . . . . . . . Sample measurement data (MS1 , Data Mode). . . . . . . . . . . . . . . . . Relative gain variations due to users in Talk Mode. . . . . . . . . . . . . . Relative gain variations due to users in Data Mode (Forward facing). . . . . Summary of relative gain over vertically polarised (VP) BS antenna. . . . . Radio signal components of a sample measurement data. . . . . . . . . . . 3.7 3.8 3.9 3.10 3.11 3.12 3.13 3.14 3.15 3.16 3.17 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 10 13 14 14 21 30 31 32 32 33 36 37 39 41 42 42 43 46 47 48 49 50 List of Figures 3.18 Summary of Rice K-factor. . . . . . . . . . . . . . . . . . . . . . . . . . . 3.19 Summary of slow fading standard deviation, σSF . . . . . . . . . . . . . . . 4.1 4.2 4.3 4.4 4.5 4.6 4.7 4.8 4.9 4.10 4.11 4.12 4.13 4.14 4.15 4.16 4.17 4.18 4.19 4.20 4.21 4.22 4.23 4.24 4.25 4.26 4.27 vii 52 54 Quadrifilar helix antenna (QHA) in the unwrapped and wrapped forms. . . 58 Scattering Parameters (S-parameters) . . . . . . . . . . . . . . . . . . . . . 60 LHCP elevation radiation gain and phase for each helical element of the QHA. 61 LHCP elevation radiation phase of each element of the quadrature-fed QHA 61 Elevation gain and axial ratio of the quadrature-fed QHA (φ = 0°). . . . . . 61 Fabricated quadrature feed network. . . . . . . . . . . . . . . . . . . . . . 62 Simulated S-parameter of the QFN. . . . . . . . . . . . . . . . . . . . . . 63 Fabricated QHA with feed networks. . . . . . . . . . . . . . . . . . . . . . 64 Measured S-parameter and radiation pattern of the reference QHA. . . . . . 65 One meandered section of the PML. . . . . . . . . . . . . . . . . . . . . . 66 PML with different number of subsection, Nl . . . . . . . . . . . . . . . . 67 PML resonant frequency. . . . . . . . . . . . . . . . . . . . . . . . . . . . 68 QHA with PML. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 69 Antenna gain over frequency with different parasitic loop (φ = 90°, θ = 0°). 70 The effect of varying Llg for QHA with PML (Nl = 16, Lmv = 5.75 mm). . . 71 Gain comparison between different parasitic loop configurations optimised for gain improvement for operating frequency between 2.35 GHz and 2.55 GHz. 72 QHA with PML(Nl = 16, Lmv = 5.8) elevation radiation pattern and axial ratio at φ =0°. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74 Fabricated PML with QHA and quadrature feed network. . . . . . . . . . . 75 Measured elevation radiation pattern of the fabricated QHA with PML (φ =90°). 76 Parasitic Quadrifilar Helix Loop (PQHL). . . . . . . . . . . . . . . . . . . 77 PQHL parameter length over frequency. . . . . . . . . . . . . . . . . . . . 78 PQHL placement on top of the QHA. . . . . . . . . . . . . . . . . . . . . 79 Antenna gain with different PQHL middle element length, Lq−middle . . . . . 79 The effect of varying Llg for QHA with PQHL (Lq−middle = 38 mm). . . . . 80 Nested PQHL. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81 QHA with PQHL(Lq−middle =38 mm, Llg =12.5 mm) elevation radiation pattern and axial ratio at φ =0°. . . . . . . . . . . . . . . . . . . . . . . . 82 Fabricated PQHL(Lq−middle =38 mm, Llg =12.5 mm) with QHA and quadrature feed network. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83 List of Figures viii 4.28 Measured elevation radiation pattern of the fabricated QHA with PQHL (2.45 GHz, φ =90°, Lq−middle =38 mm). . . . . . . . . . . . . . . . . . . . 84 4.29 SPQHE with QHA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85 4.30 Antenna radiation pattern (φ = 90°) with varying SPQHE resonant frequency, fspqhe . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87 4.31 Antenna radiation pattern (φ = 90°) with varying SPQHE and QHA separation distance, Ls−distance (Ls−middle = 83.5mm, 2.45 GHz). . . . . . . . . 88 4.32 Antenna return loss with varying SPQHE and QHA separation distance, Ls−distance (Ls−middle = 83.5mm, 2.45 GHz). . . . . . . . . . . . . . . . . . 89 4.33 Elevation radiation pattern (φ = 90°) of the SPQHE at different frequencies (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ )). . . . . . . . . . . . . . 89 4.34 Elevation radiation pattern (φ = 90°) of the open circuit SPQHE (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ ), 2.45 GHz). . . . . . . . . . . . . . . 90 4.35 Relative gain over middle radiation for the elevation radiation pattern of the open circuit SPQHE (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ ), 2.45 GHz). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 91 4.36 Fabricated SPQHE with QHA and quadrature feed network (Ls−distance =49.0 mm). 92 4.37 Measured elevation radiation pattern of the fabricated SPQHE with QHA (2.45 GHz, φ =90°, Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ )). . . 93 4.38 Optimum combining result . . . . . . . . . . . . . . . . . . . . . . . . . . 94 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8 5.9 5.10 5.11 5.12 5.13 Measurement site. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99 Base stations’ antennas . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100 SPQHA (BSB ) radiation pattern. . . . . . . . . . . . . . . . . . . . . . . . 101 Mobile stations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102 MSA return loss. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103 Radiation pattern for MSA2 . . . . . . . . . . . . . . . . . . . . . . . . . . . 103 Omnidirectional antenna, Omnirx . . . . . . . . . . . . . . . . . . . . . . . 104 Elektrobit Propsound wideband MIMO channel sounder. . . . . . . . . . . 107 Block diagram of the Elektrobit Propsound wideband MIMO channel sounder.108 Measurement paths and zones. . . . . . . . . . . . . . . . . . . . . . . . . 111 Local mean for MSA1 (MeasNear−data ). . . . . . . . . . . . . . . . . . . . . 115 CCDF plot of the local mean for MSA1 at each base station (MeasNear−data ). 116 CCDF plot of the local mean for MSA1 at each base station in measurement MeasNear−data with GAS. . . . . . . . . . . . . . . . . . . . . . . . . . . . 117 List of Figures ix 5.14 CCDF plot of the received signal for MSA1 at each base station in measurement MeasNear−data with GAS. . . . . . . . . . . . . . . . . . . . . . . . . 117 5.15 CCDF plot of the received signal for MSA1 at each base station in measurement MeasNear−data with GAS. . . . . . . . . . . . . . . . . . . . . . . . . 119 5.16 CCDF plot of the received signal for MSA1 as secondary user (MeasNear−data ).120 5.17 CCDF plot of the SIR for MSA1 (MeasNear−data ). . . . . . . . . . . . . . . 121 5.18 MIMO eigenvalues distribution with two different schemes. . . . . . . . . . 124 5.19 Capacity comparison for different MIMO schemes. . . . . . . . . . . . . . 125 5.20 A sample received signal at MSA1 from BSA−mid and BSB−mid (MeasFar−data ).126 5.21 Fast fading CDF plot for the received signal at MSA1 from BSA−mid (MeasFar−data ).127 5.22 Local mean CDF plot for the received signal at MSA1 from BSA−mid (MeasFar−data ).127 5.23 A sample of high and low regions for BSA−mid and BSB−mid with respect to MSA1 (MeasFar−data ). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 128 List of Tables 2.1 2.2 2.3 Interference scenarios in two-tier network. . . . . . . . . . . . . . . . . . . Summary of mobile phone antenna designs and their dominant polarisations in the respected positioning and modes. . . . . . . . . . . . . . . . . . . . Cross polarisation discrimination (XPD) in different environments. . . . . . 11 15 3.1 Measurement Scenarios and the number of measurement performed. . . . . 44 4.1 4.2 QHA physical parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . Physical characteristics of the PML with different number of sections, Nl and reference circular loop (Resonant frequency of 2.80 GHz). . . . . . . . Physical characteristics of the parasitic loops optimised for gain improvement for operating frequency between 2.35 GHz and 2.55 GHz. . . . . . . Physical parameters of the fabricated PML. . . . . . . . . . . . . . . . . . Fixed parameters of the PQHL. . . . . . . . . . . . . . . . . . . . . . . . . Maximum antenna gain and tilt angle, θtilt with different SPQHE middle element length, Ls−middle (2.45 GHz). . . . . . . . . . . . . . . . . . . . . Antenna gain and tilt angle with varying SPQHE and QHA separation distance, Ls−distance as a fraction of wavelength at 2.45 GHz (Ls−middle = 83.5mm, 2.45 GHz). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The SQPHE tilt angle, θtilt and the respective gain at different frequencies (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ )). . . . . . . . . . . . . . 58 4.3 4.4 4.5 4.6 4.7 4.8 5.1 5.2 5.3 5.4 5.5 Transmitters’ and receivers’ cables. . . . . . . . . . . . . . . . . . . . . . . Measurement and equipment parameters. . . . . . . . . . . . . . . . . . . Measurement scenarios. . . . . . . . . . . . . . . . . . . . . . . . . . . . . Summary of instantaneous power diversity gain in different measurements relative to the middle parasitic antenna. . . . . . . . . . . . . . . . . . . . Summary of interference scenarios at MSA 1. . . . . . . . . . . . . . . . . . 8 68 73 75 78 86 87 90 106 110 113 119 121 List of Tables 5.6 xi Markov state and transition matrices. . . . . . . . . . . . . . . . . . . . . . 129 Nomenclature Symbols α Mobile station tilt angle in the Talk Mode β Mobile station tilt angle in the Data Mode H Channel matrix hrx,tx Channel coefficient Omnirx Omnidirectional antenna Prx Receive power Ptx Transmit power Si j Scattering parameter Acronyms / Abbreviations BS Base station CCI Co-channel interference CCDF Cumulative distribution function CDF Cumulative distribution function CDMA Code division multiple access CP Circular polarisation CSG Close subscriber group CST Computer Simulation Technology Nomenclature ESPAR Electronically Steerable Parasitic Array Radiator FASPA Fixed Active Switch Parasitic Antenna FBS Femtocell base station FIT Finite Integration Technique GSM Global System for Mobile Communications HPBW Half power beamwidth HP Horizontal polarisation IMFN Independent microstrip feed network IQHA Intelligent QHA GAS Global antenna selection LAS Local antenna selection MAS Middle antenna selection LHCP Left hand circular polarisation LOS Line-of-sight LP Linear polarisation SP Slanted 45° polarisation LTE Long Term Evolution MBS Macrocell base station MIMO Multiple input multiple output MS Base station OFDMA Orthogonal Frequency-Division Multiple Access OLOS Obstructed line-of-sight OSG Open subscriber group xiii Nomenclature PCL Parasitic circular loop PDP Power delay profile PIFA Planar inverted-F antenna PL Path Loss PML Parasitic meandered loop PQHL Parasitic quadrifilar helix loop QFN Quadrature Feed Network QHA Quadrifilar helical antenna RHCP Right hand circular polarisation SASPA Switched Active Switched Parasitic Antenna SCN Small cell network SF Slow fading SIR Signal-to-interference ratio SON Self-Organising-Network SPQHA Side parasitic quadrifilar helix antenna SPQHE Side parasitic quadrifilar helix element TDM Time division multiplexing TH-CDMA Time-hopped CDMA VNA Vector network analyser VP Vertical polarisation WLAN Wireless local area network XPD Cross-polarisation discrimination xiv Chapter 1 Introduction The number of mobile phone subscriber have expanded rapidly since the introduction of digital mobile phones in the early 1990s. It is estimated to have reached 7.4 billion by the end of 2014 [1]. Traditionally, mobile phone have been used for voice calls with a data rate of only 10kb/s or less [2]. However, by the end of 2014, 55% of mobile traffic originated from high data rate internet traffic [1] such as high definition video and websites. These imply that the base station (BS) not only needs to provide high data throughput, but it also needs to be closer to users in order to ensure higher signal-to-noise ratio (SNR) with the mobile phones. A femtocell BS is a low cost and short range BS. It is originally deployed to provide coverage for small areas such as in offices and houses. However, due to its effectiveness in improving the spectral efficiency, multiple public femtocell BSs are being deployed to cover small areas in public places. A collection of public femtocell BSs in a local area is also known as a small cell network (SCN). The SCN has been shown to be a viable solution in improving the spectral efficiency to satisfy the growing demand for high data rate mobile network. 1.1 Motivation The close range public femtocells deployment creates two main challenges; firstly, the increase in cross polarization mismatch between the mobile station (MS) and the BS, and secondly, the increase in co-channel interference. In a public femtocell environment, the possibility of a line-of-sight link between the BSs and MSs is expected to be dominant due to the limited number of obstructions. The signal from the mobile station to the base station will largely maintain its original polarisation state. However, the cross polarisation mis- 1.2 Contributions 2 match between the mobile station and the base stations arises due to the random handling of the mobile station when operated in Talk and Data Modes. The deployment of multiple femtocells at close range will also increase the interference with each other as well as interference to MSs because the public femtocells will be using the same spectrum. The main objective of this thesis is to characterise the channel in a public femtocell environment and to design suitable antennas for a public femtocell. This thesis covers the evaluation of the performance BS antenna having different polarisations. Several antenna designs are also proposed. The aim of the antenna analysis is to improve the gain of the QHA and to provide the QHA with low complexity beamsteering capability. Using the QHA with beamsteering capability, a field measurement is performed to verify its effectiveness in term of diversity gain and signal-to-interference ratio improvement. A comparison to full MIMO antenna setup will also be performed. 1.2 Contributions The following are original contributions to knowledge included in this thesis: • QHA designs which yield gain improvement using a parasitic loop have been proposed. The first design is based on a parasitic meandered loop (PML) which has the same radius as the QHA. The design also helps with fabrication process as it can be fabricated on the same flexible substrate sheet as the QHA. The second design is based on parasitic quadrifilar helix loop (PQHL). Both designs have been fabricated and validated to be able to provide gain improvement. • A switched parasitic QHA (SPQHA) has been designed and its performance evaluated. By using parasitic elements at the side of the QHA, it gives QHA a low complexity beam steering capability. This feature is useful in cooperative small cell network to improve coverage and minimise interference. • Evaluation of signal polarisation effects under human handling influence in a public femtocell environment. Several BS antennas with different polarisations have been evaluated. From the analysis, it was found that polarisation mismatch between the base station and mobile can be high in such environments due to availability of lineof-sight. The analysis also has shown that a circular polarised antenna for public femtocell base station can be a good compromise compared to other polarisations for minimising the polarisation mismatch. 1.3 Publication 3 • Investigation and validation of the performance of the SPQHA in a SCN environment. The SPQHA is capable of providing high diversity gain and a good SIR improvement. Multiple Input Multiple Output (MIMO) SPQHA also has comparable performance compared to a full scale 8 element QHA-based MIMO. The analysis in this thesis has shown that MIMO SPQHA can reduce the number of RF-chain required as compared to a full MIMO setup. 1.3 Publication The following conference paper has been published: • T.F.B Tengku Mohmed Noor Izam; T.W.C. Brown, 'Evaluation of signal polarisation effects under human handling influence in public femtocell environment,' 6th European Conference on Antennas and Propagation (EUCAP), March 2012 The following journal and conference papers are being drafted for publication: • 'Cooperative small cell network with switched parasitic quadrifilar helix antennas.' Journal publication. • 'Novel parasitic loops for QHA gain improvement.' Conference publication. 1.4 Structure of the Thesis Chapter 2 of this thesis provide a review of literature on SCNs, femtocells, interferences, the effect of user handling on the performance of MSs, and existing cross polarisation discrimination measurements. Chapter 2 also provided an extensive review of literatures on the QHA covering topics such as the operation of QHA, miniaturisation techniques, bandwidth and gain improvement, multiband capability and beamsteering capability. Chapter 3 investigated the performance of several BS transmit antennas with different polarisations compared with that of the MS. It also covered the evaluation of the effect of the human body on the antenna performance in free space and real environment. A field measurement campaign was also conducted and the result are presented. The design and analysis of the QHA with parasitic element are covered in Chapter 4. First, the operation and radiation mechanism of antenna are analysed. Two parasitic loops 1.4 Structure of the Thesis 4 are designed to improve the antennas boresight gain. A third antenna design is also presented, this is based on QHA incorporating a parasitic on one side. The design provided beamsteering capability. These antennas are simulated, fabricated and validated in anechoic chamber. Finally, an evaluation of multi branch QHA is presented. Chapter 5 covers the second field measurement setup. In this chapter, the operation of the MIMO channel sounder is reviewed. The transmit and receive antennas setup are covered in detail. The chapter also described the measurement scenarios. The result of the field measurement is also covered. This chapter provides an analysis on the improvement in local mean and instantaneous power diversity gain. Finally, the performance of the SPQHA antenna are compared to QHA-based MIMO. Chapter 6 presents the conclusion of this research work and areas that can be further explored in the future. Chapter 2 Literature Review The literature review in this chapter can be divided into two main parts. The first part starts with the review on femtocell and SCN. Interference in small network will be explained along with antenna techniques for capacity improvement and interference mitigation. Next, the review considers the effect of user handling on the performance of the MS or handset. Finally, in order to understand the importance of signal polarisations in short range communication, a review on environment cross-polarisation discrimination is provided. The second part of the review covers several topics on the QHA. It provides introduction to the history of QHA development and then covers the general characteristics of the antenna and associated design improvements. These areas include miniaturisation technique, multiband capability and beamsteering capability. 2.1 2.1.1 Small Cell Network Femtocell A femtocell is a low cost and low power base station with less than 20dBm transmit power [3]. Due to limited power, the femtocell is intended to cover short range of between 10 to 50 metres radius [4, 5]. It can be deployed randomly and without prior planning by the mobile operator. It is being deployed indoor as well as outdoor such as at home, office, rural and metropolitan environments. Femtocells use internet connection as a backhaul to the core cellular communication network. Femtocells also typically share the same spectrum as the macrocells, due scarce resource of spectrum. Femtocells also have the ability to adapt to changes in the network environment automatically [6]. A key benefit of the femtocell approach is that they improve coverage, increases user 2.1 Small Cell Network 6 capacity and provides higher data rate [2]. Coverage is improved by deploying femtocells in areas where there is weak or no signal. In rural areas, even when macrocells do exist, they are deployed to cover a vast area. Average building penetration loss could account for up to 10dB to 20dB [7]. Uniform coverage cannot be guaranteed especially indoors by microcells or macrocells. In urban areas, with tall buildings, signals at ground level might be impaired by the presense of the surrounding buildings. By having an indoor femtocell, these problems can be mitigated. Femtocell technology can be deployed with any cellular network technology such as Global System for Mobile Communications (GSM), Universal Mobile Telecommunications System (UMTS), Long Term Evolution (LTE) and LTE-Advanced. In the United Kingdom (UK), LTE will occupy the 800 MHz and 2600 MHz bands [8, 9]. The 800 MHz band will occupy 791 MHz to 862 MHz (E-UTRA Band 20 [10]). The 2600 MHz band spans between 2500 MHz to 2690 MHz (E-UTRA Band 7 and 38 [10]). Other European countries generally use the same spectrum as in the UK. In the United States, LTE is generally deployed in the 700 MHz (Band 13 and 17) and 2100/1700 MHz (Band 4) bands. LTE-Advanced will be backward compatible with LTE and it is expected to occupy the same spectrum as LTE. LTE supports Multiple Input Multiple Output (MIMO) technology for up to four transmit antennas in the downlink [11]. The downlink MIMO capability is extended to support to up to eight transmit antennas in the LTE-Advanced [12]. Due to cost and space limitations, the femtocell base station might not utilise the maximum supported MIMO antenna system. The maximum femtocell transmit power for single antenna transmission is limited to 20 dBm and it is reduced to 11 dBm for 8 antennas transmission [10]. 2.1.2 Public Femtocell A SCN is a network of multiple short range base stations [13]. Each BS can transmit at low power because it only needs to provide coverage to a small local area. One example of deploying a SCN is to use multiple femtocell BSs in public places. Public places such as shopping malls, train stations, airports and open space office buildings are suitable environments for public femtocells deployment [14, 15]. These places could be characterised with large non-cluttered open space with large number of users. One suitable deployment example can be seen in Figure 2.1 that shows the London Waterloo Station during peak hours. High user density creates the demand for high data throughput. In such situation, multiple public femtocell BSs could be deployed throughput the area. Each femtocell will be covering smaller area and thus handles lower user density. 2.1 Small Cell Network 7 Figure 2.1 London Waterloo Station during peak hour (Example scenario). This ensures that the high data rate requirement can be met. The femtocells in the SCN will cooperate with each other as part of Self-Organising-Network (SON) to maximise coverage and throughput [16]. The SON will also minimise interference between femtocells and to macrocells. The environment, in which a home or small office femtocell is deployed, is very different compared to the public femtocell. Measurement in [17] has shown that a typical office environment has Rayleigh fading characteristic. Rayleigh fading indicates the absence of line-of-sight (LOS) signal component between the transmitter and receiver. A home or small office femtocell needs to cover multiple rooms separated by multiple walls. These create obstruction between the femtocell BS (FBS) and the MS. The femtocell is also typically installed at low level above the floor. As a result, the furniture and people in the surrounding can cause obstruction as well. Public places such as shopping malls, airports, railway stations and stadium typically have large open space areas. In contrast to the above situation, public places with large open space have low level of obstruction. The channel measurements in [17] and [18] have shown that large open spaces have Ricean fading characteristic with K-factor of typically more than 10dB. High K-factor indicates the existence of a strong dominant signal from the transmitter to the receiver through LOS link. This situation is often helped by installing the antenna high above the floor to further improve the LOS link. 8 2.1 Small Cell Network Table 2.1 Interference scenarios in two-tier network. 2.1.3 Co-layer interference Cross-layer interference FBS ↔ Neighbouring FBS FBS ↔ MS of a neighbouring FBS MBS ↔ FBS MBS ↔ MS of FBS FBS ↔ MS of MBS Interference in Two-tier Network A network in which there is coexistence between the femtocell and macrocell is called a two-tier or two-layer network [19]. The RF spectrum is limited and the femtocells and macrocells are typically operating in the same frequency which will give rise to co-channel interference (CCI). Interference in the network can exist in both layers as summarised in Table 2.1. Co-layer interference exists in network of the same layer such the interference between neighbouring femtocell base stations (FBSs) and the interference between one FBS with the mobile station (MS) of a neighbouring FBS. For the cross-layer, interference exists between macrocell base station (MBS) and FBS, MBS and MS of FBS as well as FBS and MS of MBS. The interference mechanism in the uplink and downlink can be different depending on the transmission technology (GSM, CDMA, OFDMA, etc.) used. Only general discussion of interference mechanism is covered in this section, without specific reference to the transmission technology used. The user access to a femtocell can be divided into three types; Closed Subscriber Group (CSG), Open Subscriber Group (OSG) and Hybrid access mode [20]. Only authorised users will able to access a particular femtocell in CSG mode while all users will have access to the femtocell in OSG mode. Hybrid access mode will give higher priority access to certain users and lower priority access to other users. In CSG mode, the FBS could cause interference to the MSs of the neighbouring FBSs and also to the MSs of the MBSs. Similarly, the MSs of the MBSs could also cause interference to the FBSs close to it. This problem is known as ‘Loud Neighbour’ [21] or “Near-far” problem [19]. It could be mitigated through power control [22, 23] or adaptive power level [2, 24] on the FBSs, MBSs and MSs. This problem could also be mitigated by allowing the MS to connect to the closest base station with lower transmit power in OSG access mode [19]. One of the simplest ways to eliminate interference is by assigning different frequencies for FBSs and MSs. However, this method is not practical due to limited spectrum available to each mobile communication carrier. In order to resolve this, the limited spectrum available to each carrier can be further divided in smaller sub-channels to be used among 2.1 Small Cell Network 9 femtocells and macrocells through spectrum partitioning or splitting [19]. One such example is the frequency hopping in GSM network [2]. This will eliminate interference as the BSs operate at different sub frequencies. In an OFDM system, the spectrum already consists of several sub-channels [20]. FBS and MBS will be using different sub channels groups as determined by the Fractional Frequency Reuse (FFR) factor [25] and thus eliminating the interference [4]. A drawback of spectrum partitioning is the lower overall spectral efficiency of the network. Instead of sharing the spectrum by splitting in the frequency domain, the transmission can also be partitioned in the time domain. In time-hopped CDMA (THCDMA) systems, the transmission period is divided in to several slots and the femtocell will only transmit in one of these time slots [2]. Interference will be reduced but there is also loss in processing gain [4]. 2.1.4 Antenna Techniques for Capacity Improvement and Interference Mitigation Antenna sectorisation employs multiple antenna elements arranged to provide coverage in different directions, which will help in improving coverage and minimising the interference. Due to cost limitation, the base station can use a single RF chain and coverage is changed by switching between the sectorised antenna elements. This method has been found to be effective, as evaluated in [4, 26]. With the usage of antenna sectorisation with directional antenna, the capacity and interference can be further improved for a femtocell base station. Directional antenna is preferable to omnidirectional antenna due to its low side-lobe and high boresight gain. A low side-lobe helps in reducing the interference from adjacent femtocells while higher antenna gain at the boresight improves the signal-to-noise ratio (SNR) and link reliability. Evaluation in [26, 27] has shown that a 5dBi directional horn antenna has 58% higher femtocell capacity than a 3dB omnidirectional antenna. Using antenna selection technique, the femtocell base station is also equipped with multiple antennas. The system can have the same number of RF chains as the number of antennas or lower number of RF chain to reduce cost. The subset of antenna with highest SNR will be chosen during operation which will provide diversity gain to the system. In [28], a 1 dB to 2 dB diversity gain is obtained by using this technique with four antennas with one RF chain. For a higher number of antennas and RF chains, interference rejection combining (IRC) could be used to reduce the complexity of the antenna selection technique [29]. For a more complex system with multiple RF chains, beamforming technique could be 10 2.2 Operator’s Influence On Mobile Phone Antenna (a) Talk mode (b) Data mode Figure 2.2 Mobile station orientations. employed to further reduce interference and improve coverage. Using this technique, the gain and phase of each antenna element is manipulated before being combined in order to direct the radiation towards a certain direction. In [30], using this technique with six antenna elements has been found to improve coverage and reduce the pilot signal power which in turn reduces the interference. By using two planar inverted-F antennas and two patch antennas with smart configuration, evaluation in [31] has shown that an average of 2.5 dB signal gain compared to omnidirectional antenna can be achieved. However, with the increased complexity of multiple RF chains, cost will rise. Another antenna technique that could be used for femtocell base station is distributed antenna system (DAS). The DAS utilises multiple antennas but these antennas are installed in various part of a building to provide full coverage. Instead of having a complete RF chain at each antenna site, these spatially separated antennas are connected to a common femtocell base station for a joint signal processing [32–34]. These antenna needs to provide coverage to a smaller area and can be switched off when there is no demand. With lower power, interference can also be reduced. The spatially separated antennas will also provide diversity to improve reliability. One of the drawbacks of a DAS femtocell is that it needs a careful RF planning when installing the antennas. 2.2 Operator’s Influence On Mobile Phone Antenna The internal antenna has to be designed to an even smaller size nowadays as the mobile phones are getting smaller in size. This will deteriorate the performance as an electrically 11 2.2 Operator’s Influence On Mobile Phone Antenna Table 2.2 Summary of mobile phone antenna designs and their dominant polarisations in the respected positioning and modes. Antenna Upright Data Mode (Portrait) Data Mode (Landscape) Reference Shorted Patch Printed Monopole slot Coupled-fed PIFA Planar monopole Folded planar monopole Planar monopole Crossed C-shaped VP VP VP/HP VP VP VP VP/HP VP HP HP/HP VP VP/HP VP/HP HP HP HP HP/HP HP HP HP HP [42] [43] [44] [45] [46] [47] [48] small antenna is also more susceptible to objects in its surrounding [35], known as near field effects. This is because, when operated by human, the near field of the antenna will be in close proximity with human hand and head [36]. The human tissue can be considered as lossy dielectric material [37] and up to 10 dB of performance loss could be experienced when both head and hand are present [38]. The shoulder also acts as a poor reflector and absorbs part of the radiation from the antenna [37]. The complex interaction between the antenna and human tissue could detune the resonant frequency of the antenna which will lead to impedance mismatch loss [39]. Due to absorption, reflection and diffraction, the head and hand could also change the radiation pattern and dominant polarisation of the antenna compared to its free-space radiation pattern [38, 39]. Nowadays, the mobile phone is no longer mainly used for making calls (Talk Mode), but also for watching video and surfing the Internet (Data Mode). The user will handle the mobile phone differently in these two situations, as shown in Figure 2.2(a) (Talk Mode) and Figure 2.2(b) (Data Mode) [36]. In Talk Mode, the mobile phone is inclined at angle α from the vertical position and held by the side of the head. In Data Mode, the mobile phone is inclined at angle β from the horizontal position and held in front of the body. For the Data Mode, the mobile phone can be used in portrait or landscape orientations. The inclination angle, α and β varies from user to user. Based on previous studies in [40, 41], the average inclination angle in Talk Mode, α has been found to be about 60°. BS antennas are traditionally designed to be vertically polarised and due to that, the mobile phone antenna is designed to be vertically polarised. Several commonly used mobile phone antenna designs are summarised in Table 2.2. It can be seen in the table that, depending on the mobile phone antenna design, the dominant polarisation could switch through 90° between Talk and Data Modes (portrait or landscape); causing polarisation mismatch 2.3 Signal Polarisation and Cross-polarisation discrimination (XPD) 12 loss. Under daily usage, antenna misalignment between the BS and mobile phone due to human handling in these two modes can not be avoided. A polarisation mismatch loss of up to 10-20 dB when the two antennas are not perfectly aligned [49] could be experienced especially for handsets in Data Mode. The position of fingers when holding the mobile phone could also deteriorate the antenna performance. An extensive study of the effect of hand grip style and finger positions for different mobile and antenna designs can be found in [36, 40]. 2.3 Signal Polarisation and Cross-polarisation discrimination (XPD) 2.3.1 Signal Polarisation in Free Space Polarisation mismatch loss occurs when the polarisation of the transmitted signal is not matched with the polarisation of the received antenna. Polarisation efficiency, PE or polarisation loss factor (PLF) is defined as [50, 51]: ∗ 2 PE = |ρin • ρant | (2.1) where ρin and ρant are the polarisation unit vectors of the incident wave and the receiving antenna, respectively. Polarisation efficiency of 1 indicates no polarisation mismatch loss while polarisation efficiency of 0 indicates total loss. The polarisation unit vector for linear, right hand circular and left hand circular polarisation are given by Eq. (2.2), Eq. (2.3) and Eq. (2.4), respectively [51]. For the linear polarisation, θ is the slant angle of the polarisation unit vector with respect to vertically aligned y-axis with wave propagation in the +z-direction. Circular polarised signals can be regarded as the combination of a vertically and linearly polarised signal with 90° phase difference between them. ρLP = cos θ xˆ + sin θ yˆ (2.2) 1 ˆ ρRH = √ (xˆ − jy) 2 (2.3) 2.3 Signal Polarisation and Cross-polarisation discrimination (XPD) 13 Figure 2.3 Separation angle in linearly polarised transmission systems. 1 ρLH = √ (xˆ + jy) ˆ 2 (2.4) When both the incident wave and the receiving antenna are linearly polarised, the polarisation efficiency simplifies to PE = | cos γ|2 (2.5) where γ is the separation angle between the polarisation unit vectors of the incident wave and the receiving antenna (Figure 2.3). The polarisation efficiency for this combination is plotted in Figure 2.4. When they are perfectly aligned (γ = 0°), polarisation efficiency is equal to 1. Polarisation efficiency reduces to 0 when the polarisation of the incident wave is orthogonal (i.e. γ = 90°) to the polarisation of the receiving antenna. At 45° tilt angle, polarisation efficiency is 0.5 or −3 dB. When a circular polarised wave is received by linear polarised antenna or vice versa, it can be shown that polarisation efficiency is provided by 1 PE = (cos2 θ + sin2 θ ) 2 (2.6) The derivation of Eq. (2.6) can be found in Appendix A. The term in the parentheses on the right hand side of Eq. (2.6) is always equals to unity for any slant angle, θ . The polarisation efficiency is always equal to 0.5 or −3 dB irrespective of the slant angle of the linear polarised signal. 2.3 Signal Polarisation and Cross-polarisation discrimination (XPD) (a) Linear 14 (b) Decibel(dB) Figure 2.4 Polarisation efficiency vs. tilt angle for linear polarised antennas. Figure 2.5 Signal depolarisation in a real environment. 2.3.2 Cross-Polarisation Discrimination (XPD) In a cluttered environment, signal from one polarisation state could couple into the orthogonal polarisation state due to reflection, diffraction and scattering. For example, when a vertically polarised signal is transmitted, part of this signal will get depolarised into the horizontal polarisation during propagation to the receiver. At the receiver, both vertically and horizontally polarised signal will be available. This is illustrated in Figure 2.5, for a vertically and horizontally polarised transmitted signal. The level of signal depolarisation is characterised by cross polarisation discrimination (XPD) as the ratio of co-polarised received power to the cross-polarised received power. The XPD for vertically and horizontally polarised transmitted signal is given by Eq. (2.7) and (2.8), respectively. The first and second subscripts, on the right-hand-side of the equations, 15 2.3 Signal Polarisation and Cross-polarisation discrimination (XPD) Table 2.3 Cross polarisation discrimination (XPD) in different environments. Indoor Outdoor XPDV (dB) 12 7 6 OLOS 6 15 LOS [email protected] 17 0 2.5 OLOS [email protected] m 8.6 LOS Freq. (GHz) Dist. 0.463 0.463 0.836 0.92 1.95 2.6 2.4 1.95 0.8 2.6 2.4 20 km Suburban 20 km Urban >3 km Suburban >2 km Metropolitan area 1-80 m Hallway (6m wide) 2-50 m Office building corridor 1-14 m Corridor 1-80 m Laboratory 12x24x10 Inside house 2-50 m Office building corridor 1-14 m Corridor Site Reference [54] [54] [55] [56] [57]] [58] [53] [57] [59] [58] [53] refer to the polarisation of the receiving and transmitting and antenna, respectively. XPDV = PVV PHV (2.7) XPDH = PHH PV H (2.8) The XPD level, as measured in various environments are summarised in Table 2.3. The list is not comprehensive but it gives the an overview of different XPD levels for different scenarios. Factors such as environment, frequency, and distance could affect the measured XPD level. In many literature sources, the difference between the XPDV and XPDH is rarely reported. This is either because they are difficult to measure or the difference is negligible. The difference between indoor NLOS XPDV and XPDH is only 0.1 dB [52]. In [53], the difference is reported to be 2.5 dB in LOS and 0.3 dB in NLOS. The level of XPD differs significantly depending on the link between the transmitter and receiver. In a line-of-sight (LOS) environment, the transmitted signal propagates to the receiver through a direct route and largely maintains its original polarisation state. The level of depolarisation by the environment is low. It can be seen in Table 2.3 that, the XPD in LOS could be as high as 17 dB. In non-line-of-sight (NLOS) environment, any obstruction between the transmitter and receiver will caused the signal to propagate through an indirect route which will increase the level of signal depolarisation, thus lowering XPD. It can also 2.4 Base Station Antenna 16 be seen, in Table 2.3 that the XPD for NLOS is generally lower than the LOS and could be as low as 0 dB. While many literature sources only reports the average XPD for a particular environment, the XPD is actually varies with distance. Empirical studies of the dependency of XPD on propagation distance for outdoor environment can be found in [56, 60, 61]. In the outdoor environment, measurements show that the XPD decreases with increasing distance. However, the decay rate of XPD over distance in the outdoor environment is low and the XPD is roughly constant over a short distance. In contrast to the outdoor environment, the XPD for the indoor environment increases with increasing distance as reported by [57, 58]. The XPD in LOS case rises faster compared to NLOS. The empirical relationship between XPD and distance for LOS and NLOS indoor environments is provided in [58]. In a short range open space communication link, it is expected that LOS link between the BS and MS is usually available. Even when there is an obstruction between BS and MS, a dominant reflected signal will still reach the MS. Under these situations, it is expected that the XPD will remain high. However, it is also dependent on the mobile phone antenna and its orientation. 2.4 2.4.1 Base Station Antenna Antenna Elements Wire monopole and dipole antennas are two of the simplest and most commonly used types of antenna. They are suitable for ceiling mounted base stations as they have an omnidirectional radiation pattern. Wire monopole and dipole antennas typically suffer from narrow bandwidth but this can be improved tremendously by adopting the planar form [62– 64]. Several bandwidth improvement techniques, such as bevelling and shorting the planar monopole are discussed in [65]. Monopole or dipole antennas could also be printed onto substrate to produce printed planar antennas for a more compact form factor [66]. Patch antenna are also attractive for use in BS due to their low cost and ease of fabrication. The patch antenna is also light weight, compact and suitable for low profile applications. It is intended for narrowband and low power applications and suffers from high ohmic loss, low efficiency and low polarization purity [67, 68]. Backward radiation could also be a problem for a patch antenna with small ground plane. The bandwidth can be widened by incorporating design improvements such as slits [69] and slot [70]. Other design improvements to improve the performance of patches antenna can be found in [68, 71]. 2.4 Base Station Antenna 17 A reflector antenna is typically used in applications that need high gain antennas. They consist of radiating elements and a reflector plane. The reflector plane comes in various shapes such as flat, corner and parabolic. The reflector plane helps to redirect or concentrate the radiation towards the desired direction. For very high gain applications, a parabolic reflector is typically used and the antenna’s gain is directly proportional to the square of the parabolic reflector’s diameter [72]. The radiating element is typically placed at the focal point of a parabolic reflector or a quarter wavelength from a flat reflector plane [73]. This requirement will result in a large antenna. The use of electronic bandgap material, FabryPerot or partially reflecting surface to replace flat reflector have been shown in [73–75] to be capable of reducing the spacing between the antenna and the reflector. Most linear antennas can be oriented to obtain either vertical, horizontal and slanted polarisations. When two antennas are co-located and orthogonally fed or oriented such as in crossed-dipole and crossed-patches antennas, respectively, dual-polarised antenna can be obtained which is useful for achieving polarisation diversity gain [76]. When certain patch designs prevent the orthogonal patches from being co-located on the same plane, a stacked patches configuration could be used to achieve similar result. In order to obtain a higher gain base station antenna, several antenna elements can be combined in an linear array form [77] The dual-polarised antenna discussed above could also radiate circular polarised (CP) signal when fed with equal power but with quadrature phase difference between the two feeds. Another method to obtain a CP signal from a square patch antenna is by truncating the two diagonal corners of a square patch antenna [78]. A helical antenna with diameter of more than one wavelength is capable of radiating a CP signal [72]. However, due to its size, it might be not be the best choice as FBS antenna. An outdoor BS antenna such as for macrocell has typically high gain and placed 30 m to 70 m above the ground such as on a mast or on top of building [79]. In order to increase frequency reuse factor, a BS is usually divided into three sector or more. Each sector has its own antenna in order to provide omnidirectional coverage around the base station. The polarisation of macrocell BS is typically vertically polarised or dual polarised with ±45° polarisation [77]. A macrocell antenna is also typically equipped with downtilt capability. This can either be achieved through mechanical or electrical mean. The ability to adjust downtilt is needed in order to adjust coverage and to reduce interference to another cell [80]. An indoor BS antenna for picocell or femtocell are typically wall or ceiling mounted. This is to minimise shadowing by human and furniture. A ceiling mounted antenna usually has omnidirectional radiation pattern such as the one designed in [81]. This antenna is based 2.4 Base Station Antenna 18 on monopole antenna which radiate vertical polarisation signal. A wall mounted BS antenna is usually vertically polarised or slanted 45° polarised [82]. The QHA is one of the best candidates for CP antenna due its ability to radiate a CP signal over a wider beamwidth with excellent axial ratio [83]. The QHA consists four elements which are wrapped around an axis or a central core to form equally spaced helices on the circumference of the central core. The elements are fed with the same power at 90º phase difference. Due to its balanced configuration, it is less susceptible to ground plane size or objects in its proximity [72, 83]. This antenna will be reviewed in detail in the next section. 2.4.2 Reconfigurable Antenna Reconfigurable antenna is a form of smart antenna which has the capability to adapt its operating and radiation characteristics in order to improve its performance. This is usually achieved by activating or deactivating element or switches on the antenna to change its characteristic. The antenna in [84] is able to reconfigure it operating frequency by varying the bias voltage of the varactor diodes that have been strategically placed on the antenna. The reconfiguration of antenna polarisation is also possible. In [85], the antenna is able to change its circular polarisation sense from right-handed to left-handed. The ability to change polarisation between horizontal, vertical and 45° polarisation has been demonstrated in [86]. Another characteristic that needs reconfiguration ability is radiation pattern. The antenna in [87] and [88] have the ability to change it radiation pattern to three and eight different patterns or directions, respectively. This is important so that the antenna can maximise its coverage to the intended users or minimise interference to unintended users. 2.4.3 Parastic Antenna In general, parasitic antenna is consists of one driven element or antenna and one or more parasitic elements. When the driven antenna is radiating close to a near-resonant parasitic element, the energy from the driven antenna will couple to the parasitic element [89]. The parasitic element will then reradiate and the resulting radiation will be away from the parasitic element [90]. The parasitic elements can be of any shape provided that its resonant frequency is close to that of the driven antenna. A classic example of the application of parasitic elements is to improve the gain as can be seen in the Yagi-Uda antenna. The Yagi-Uda consists of one driven antenna with one parasitic reflector element and several parasitic director elements [89, 91]. When the 2.4 Base Station Antenna 19 resonant frequency of the parasitic element is slightly lower than that of the driven antenna, the induced current will be lagging in phase and the parasitic elements will act as a reflector [89]. On the other hand, when the resonant frequency of the parasitic element is slightly higher than the driven antenna, the induced current will be leading in phase and it will act as a director. The other resonant length, the parasitic element will be transparent to the driven antenna. The role of the parasitic elements either as a director or reflector could be changed by incorporating a switch on the parasitic element to change its resonant length [92]. In order to control the direction of radiation, the parasitic antennas can be configured in Switched Active Switched Parasitic Antenna (SASPA) or Fixed Active Switch Parasitic Antenna (FASPA) mode [90]. In SASPA configuration, the driven antenna and parasitic element(s) has the same element design. They could switch role such that the location of the driven antenna relative to the parasitic elements could be changed by using switches such as RF and MEMs switches [93, 94]. Switching loss will occur due to the switching between driven antenna and parasitic element(s). The FASPA configuration will avoid this by having a fixed driven antenna. Switching only occurs on the parasitic element(s) surrounding the driven antenna. In order to avoid variation in the input impedance and radiation pattern, the antenna and parasitic elements symmetry between switching configurations needs to be maintained [95]. By using switches, the parasitic elements could only be switched on or off resulting in fixed or sectored directions [94, 96]. The number of available radiation direction is determined by the number of switching combination of the parasitic antennas. In order to achieve higher degree of freedom, the parasitic elements could be reactively loaded [97]. Reactive loading on the parasitic elements will introduce delay during re-radiation. As a result, the tilt angle or steer direction of the parasitic antenna could be variably changed to create an Electronically Steerable Parasitic Array Radiator (ESPAR) antenna [98, 99]. With that capability, adaptive beamforming to steer the beam or form null is possible [96]. For direction finding, direction-of-arrival (DOA) algorithms such as MUSIC or ESPIRIT could be used [100, 101]. A parasitic antenna could also provide diversity gain through pattern diversity [102, 103]. An example of a patented ESPAR antenna design can be found in [104]. The antenna radiates with vertical polarisation. It is constructed using one radiating dipole in the middle with six parasitic dipole uniformly placed around the radiating element. The parasitic elements is equipped with variable reactance element. The antenna gain in the direction of the parasitic element by varying the reactance element. By configuring the parasitic element to be director or reflector, the radiation pattern of the ESPAR antenna can be changed. 2.5 Quadrifilar Helix Antenna (QHA) 20 An example of planar parasitic antenna can be found in [105]. This antenna is based on printed patch antenna. It can radiate with linear or circular polarisations depending on the input configuration. At each side of the radiating patch, there are two parasitic patch and this antenna is capable of tilting the radiation in the azimuth and elevation direction. This antenna utilises two parasitic patch at each so as to improve the maximum tilt angle and it is capable of tilting the antenna the antenna by up to approximately 35°. This antenna is designed for operating frequency of 4 GHz. Although the antenna is in planar configuration, its width is approximately 14 cm. If this antenna is to be scaled for operating frequency at 2.4 GHz, it is expected to be much larger. Each of the parasitic elements has switches in order to control their parasitic behaviour such as to turn it on or off. 2.5 2.5.1 Quadrifilar Helix Antenna (QHA) Introduction The QHA has great merits compared other circular polarised antennas such as crossed square patch, truncated square patch and helical antennas. This is due to its ability to produce a high quality circularly polarised radiation pattern over a wider beamwidth with excellent axial ratio. The QHA can also be designed to be compact and low profile as will be discussed in a later section. The multiple turn helices with phase quadrature feed QHA was invented by Gerst and Worden in 1966 [106]. It was then extensively studied by many researchers as can be found in [107–111]. Based on the work of Wheeler in [112], the fractional turn winding resonant QHA with cardioid-shaped radiation pattern was introduced by Kilgus [108, 113]. The cardioid-shaped radiation pattern is obtained when the QHA is fed in quadrature phase [113] instead of in parallel as done by Wheeler. The QHA is based on two bifilar helices with turn winding of less than one and helices element lengths of less than or equal to one wavelength (λ /4, λ /2, λ /3 and λ ). In general, the QHA consists of four elements which are fabricated from thin wires or printed on a flexible dielectric film [83]. It is then wrapped around an axis or a central core with radius, r to form equally spaced helices on the circumference of the central core. Figure 2.6 shows the printed QHA in unwrapped and wrapped forms where the numbers and α indicates the feed sequence and pitch angle, respectively. The non-feeding ends are openended when the element length, Lelem is one quarter or three quarter wavelength. When the element length is half or one wavelength, the non-feed ends are short-circuited to each other [114]. 21 2.5 Quadrifilar Helix Antenna (QHA) (a) In unwrapped form (b) In wrapped form Figure 2.6 Printed quadrifilar helix antenna (QHA). The central core is typically in the shape of cylinder but it can also assume other shapes such as cone [115] and square [116]. The central core only acts as a supporting structure and can be removed if necessary. The axial length, Lax is related to the number of turn, N, radius of the central core, r, and length of helix elements, Lelem by r Lax = N 1 (Lelem − Ar)2 − (2πr)2 2 N (2.9) where A is 2 for Lelem of half or one wavelength and 1 for Lelem of one or three quarter wavelength [109]. QHA with larger multiple of quarter wavelength element length has higher boresight gain [117]. If the element pitch angle and radius of the QHA is fixed, longer elements have to wrap around the central core which in turn increase the number of turns. Higher number of turns will yield higher boresight gain. The change in element length and number of turn will also affect the antenna’s physical dimensions, efficiency and impedance [118]. The helices’ winding direction and quadrature feed sequence will determine the sense of circular polarisation and the direction of the main beam. Right-hand circular polarisation radiation can be obtained by winding the helices in the left-hand screw direction. For forward radiation, the antenna is fed with equal power and phase of −90°, 180°, 90°, 0° for port 1 through port 4. The polarisation of the antenna will not change if the quadrature phase feed sequence is reversed. However, the antenna will radiate in the opposite direc- 2.5 Quadrifilar Helix Antenna (QHA) 22 tion [113]. In order to achieve right-hand circular polarisation in the forward direction, the winding direction and quadrature phase feed sequence are reversed. 2.5.2 Miniaturisation Techniques The axial length or height of the QHA could be a disadvantage compared to a planar antenna. In order to meet the requirement for small-form factor FBS, the QHA needs to be miniaturised. Several methods such as variable pitch angle, meandering, folding, and dielectric loading have been employed to reduce the axial length, Lax of the QHA. A combination of these techniques can also be employed to further reduce the size of the QHA. The variable pitch technique is capable of reducing the axial length by about 10% [119– 121]. Using a variable pitch angle technique, the helix elements of the QHA were divided into several sub segments and each sub segment was tilted at different pitch angle. The pitch angle has no significant effect on the input impedance but it will change the radiation pattern compared to a constant pitch QHA [119]. In order to obtain the same radiation pattern, optimisation techniques such as simulated annealing (SA) can be employed to find the optimum pitch angles for each sub elements as discussed in [119] where the pitch angles were varied between 30° and 68°. In [121], regressive pitch angle technique was used where the pitch angles were reduced at a constant step from the feed to the other end. Rectangular [83, 122], triangular, and sinusoidal [114] profiles have been used to shorten antenna height. By bending the antenna element, the mutual capacitance in each meandered section increases [123]. Since the meandered sections are joined in series, the total capacitance decreases. For the same antenna height, meandering line technique requires longer antenna element due to the bending which in turn increases the total inductance [123, 124]. The increase in total inductance dominates over the decrease in total capacitance with the net effect of lowering the resonant frequency and as a result, antenna height can be shortened in order to obtain the original resonant frequency. In [83], meandering technique with a square profile was used to reduce the axial length by 53%. A combination of sinusoidal profiles was designed in [114] to reduce the axial length by 72%. Using a similar concept to meander line, the axial length of the QHA could also be reduced by folding the helix elements. Instead of folding the elements in small meandered section, the element is folded or bent back. In [125], an axial length reduction of 20% was achieved by bending a portion of the non-fed ends of the helix elements without any detrimental impact on the antenna performance. By folding the helix elements in rectangular spiral form, the axial length was reduced by 43% as can be seen in [126, 127]. For a simple 2.5 Quadrifilar Helix Antenna (QHA) 23 single fold and double fold configuration, a reduction of 60% and 70%, respectively were achieved in [128]. The velocity of wave propagation is inversely proportional to the square root of the relative permittivity, εr of the propagation medium. The wavelength is decreased in high relative permittivity medium. Using this concept, dielectrically-loaded central core has been used to reduce the axial length of the QHA [129–131]. In [130], the central core had a relative permittivity of 40 allowing the axial length to be reduced by more than 80% compared to the reference antenna in [126]. A parametric study on a dielectrically-loaded QHA can be found in [132]. Among the disadvantages of this technique are increase in weight and changes to the input impedance of the antenna. A combination of the aforementioned miniaturisation techniques could also be employed to reduce the axial length as well as to improve the performance of the QHA. A combination of variable pitch and meander line technique has been employed in [121] to achieve 55% axial length reduction. This is a further 10% reduction compared to the meander line technique only. By combining meander line and folded techniques, the axial length reduction increases to 60% [133] from 43% [126, 127]. 2.5.3 Bandwidth and Gain Improvement The QHA typically has small bandwidth of around 5% to 8% [120]. The bandwidth depends primarily on the geometry of the QHA [134] which will affect the impedance matching with the feed network. In [101], the studies showed that the bandwidth increases with the increase in the ratio of diameter to axial length of the QHA. The bandwidth also depends on the width of the printed helix elements. Instead of using a constant width, tapered helix elements with decreasing width was proposed in [120] to increase the bandwidth from 5% to 16%. A parasitic helical strip has also been found to help in improving the bandwidth of the QHA [135, 136] by cancelling the susceptance of the radiating helix elements. This technique has been able to improve the bandwidth of the QHA to 30% at 1.5GHz. In general, the gain of the QHA can be increased by increasing the number of turns, N of the helix elements. However, for short helices, as the number of turn increases, the spacing between the helices will decrease. This will give rise to mutual coupling between the elements which is not desired. One method to control gain that does not involve in the change of antenna geometry is by introducing parasitic loop at a certain distance above the QHA [137]. The parasitic loop could be designed to act as a director to improve gain or reflector to reflect the radiation to the opposite direction. In [138], a parasitic cross dipole 2.5 Quadrifilar Helix Antenna (QHA) 24 is used to improve the gain of the QHA. Both designs increase the radius of the antenna structure. 2.5.4 Multiband Capability Apart from having a wide bandwidth, the QHA also needs to have multiband capability in order to operate in various bands for different cellular technologies such as 3G and LTE. The multiband capability could be achieved by using one of the three general techniques; discrete QHAs, band-switched elements, or multiband element designs. The first technique to achieve multiband capability is by having two discrete QHAs which operate at different frequencies. The QHAs could be arranged in vertical stack configuration [109, 139–142]. This configuration has the drawback that it increase the height of the overall structure. The top QHA could also affect the radiation pattern of the bottom QHA. In order to eliminate these drawbacks, the second QHA could be placed inside the first QHA [109, 139, 142, 143]. However, due to close proximity of the QHAs, their performance could be adversely affected. Instead of feeding the two discrete QHAs, the inner and outer QHAs could also be fed in parallel and dual band could be achieved based on the mutual coupling effect between the inner and outer elements [144]. Using octafilar configuration, each set of four alternate helix elements can be designed to resonate at two set of frequencies to essentially create two discrete QHAs [139, 145]. Mutual coupling could start to affect the antenna performance as the spacing between elements is reduced. Depending on space limitations and band separation frequency, discrete QHA technique could limit the band to only two frequency bands. As the resonant frequency of the QHA depends on the length of the helix elements, Lelem , multiband operation could be achieved by changing the element lengths during operation. A pin diode was used in [146] to act as a switch to either shorten or lengthen the helix elements. In order to avoid switching, lumped components could also be used to act as a filter [147]. The lumped elements, consisting of parallel capacitor and inductor were attached in the helix elements and it will be open circuit at its resonant frequency. This will isolate the top portion from the bottom portion of the QHA to effectively change the QHA’s resonant frequency. Design improvement to the helix elements could also help in obtaining multiple band capability. In [126, 127], the elements are folded into a rectangular spiral which resonate at three different frequencies. Using a meander line and folding technique, dual band capability is obtained where the lower band depends on the length of the folded section [127, 133]. 2.5 Quadrifilar Helix Antenna (QHA) 25 Another design improvement that can be incorporated to the helix elements is the multiple arm technique [148, 149]. Using this technique, several slots are introduced to the non-fed ends of the helix elements to create several thin arms with different length and as a result, the QHA resonate at three different frequencies. 2.5.5 Intelligent QHA (IQHA) The IQHA, invented at the University of Surrey has the capability to adapt to various operating conditions by intelligently adjusting its operating frequency bands, radiation modes, feed matching circuit, and feed weighting [150, 151]. In contrast to conventional quadraturephase fed QHA, the signal from each element of the IQHA could be combined with different weights to provide diversity gain. Due to the small spacing between elements, the diversity gain is primarily achieved from the angular diversity [152]. Using maximal ratio combining (MRC), the magnitude and phase of each signal could be combined accordingly. Equal gain combining (EGC) is simpler than MRC where only the phase component of each signal is adapted and this combiner has been shown to provide up to 14 dB of diversity gain in NLOS environments [153]. However, the diversity gain diminishes in LOS environment where the correlation between elements is higher. The IQHA is also a good candidate for use in MIMO systems where it would reduce the space occupied by the array antennas. Compared to an array of four dipole antennas with half wavelength separation distance, a MIMO IQHA provides slightly less capacity but with 70% reduction in size [118]. A linear or rectangular array of 8 antennas is not practical especially for FBSs and this can be replaced with two spatially separated IQHAs. In [154], a 8x4 MIMO channel measurement based on two QHAs on one end has been evaluated on which only four out of eight elements of the IQHAs are selected at one time. The antenna selection technique enables the capacity improvement by up to 30% over a single MIMO IQHA in NLOS environment and as expected, the capacity improvement is lower in LOS environment. 2.5.6 Beamsteering Capability A femtocell base station needs the ability to steer the radiation beam towards a certain direction. This is important primarily for two reasons. First, the femtocell base station needs to be able to adapt its coverage area in the case new femtocells are deployed nearby in order to minimise interference among the neighbouring femtocells. Secondly, a group of neighbouring femtocell could be intelligently configured to serve different cluster of users by radiating 2.6 Summary 26 mainly towards that clusters. This, in turn will help in maximising the throughput among the users. Beamsteering Capability based on IQHA The IQHA is capable of changing its radiation pattern shape by adjusting the relative phase difference between the helix elements. While a conventional phase-quadrature fed IQHA will radiate in cardioid pattern, an IQHA when fed with either equal phase or 180º phases different between elements will radiate with a monopole-like radiation pattern [151]. This is useful to provide greater coverage but at the expense of lower gain and losing the circular polarisation radiation. In general, IQHA has a limited beamsteering capability. In [155], it has been shown that beam steering for IQHA is possible to a limited extent by varying the relative phase between elements at certain configurations. Although the purpose was to steer the beam away from mobile users’ head, it is also applicable for use in FBS to tilt the beam towards certain directions. There is up to 2.9 dB beam steering gain but this is at 3 dB gain loss compared to the gain with phase-quadrature feed in the boresight direction. One of the aspects that was not evaluated is the impact on the axial ratio. Axial ratio is important in determining the quality of the circular polarisation wave. Beamsteering Capability based on Parasitic QHA The application of a parasitic element in a QHA could provide a simple and low cost solution to obtain beamsteering capability. Using parasitic elements, only one RF chain is needed when the QHA is fed in phase-quadrature compared to four RF chain in IQHA. However, the parasitic QHA has not been thoroughly studied. To date, the only reference that could be found is a conference paper in [156]. It is based on simulation of 2 by 2 QHA array spaced at 0.2λ in SASPA configuration. It was able to achieve tilt angle of 25º with reasonable gain. The effect of separation distance on antenna performance and its full potential has not been discussed. 2.6 Summary The femtocell BS has emerged as a cheap and viable solution for improving coverage, increasing capacity, and enhancing data throughput. Although the femtocell was originally 2.6 Summary 27 designed as a short range indoor base station to be used at home or small office, the femtocells are now being deployed to provide network coverage in public places with high user density such as shopping malls and stadiums. Multiple femtocells deployment in large open space public places will form a SCN, in which he node can form a self-organising-network (SON). They will cooperate among themselves to minimise inter-cell interference while at the same time trying to maximise the data throughput to cover the high number of users. The environment, in which a home or small office femtocell is deployed, is very different compared to the public femtocell. A home or small office femtocell needs to cover multiple rooms and the multiple walls create obstruction between the femtocell BS and the MS. The femtocell is also typically installed low above the floor. The furniture and people in that place can cause obstruction. Due to the rich obstructions, the signal transmitted by the femtocell base station gets depolarised before reaching the mobile station and as a result, the transmitted signal does not maintain the original polarisation state. In contrast to above situation, a public femtocell is typically deployed to cover large open space and installed high above the ground and as a consequent, obstruction due to the walls or objects in the surrounding is limited. Even when there are obstructions between the base station and mobile station, a dominant reflected signal will still reach the mobile station. The transmitted signal from the femtocell base station will largely maintain its original polarisation state when reaching the mobile station. Various existing cross-polarisation-discrimination (XPD) measurements agree with this assumption. The XPD value in a short range LOS environment has been shown to be high which indicates that the transmitted signal maintains its original polarisation state. Even though the depolarisation effect is low in the public femtocell environment, the polarisation mismatch loss between the femtocell BS and MS can still be high. Another factor that gives rise to the polarisation mismatch loss is human handling. When held close to human body, the performance of the antenna can deteriorate and the dominant polarisation can change. The MS is also held differently in Talk and Data Modes. When coupled with different antenna design and placement in the MS, the MS antenna polarisation with respect to the base station can be said to be random. This causes the polarisation mismatch between the femtocell BS and the MS. This effect needs to be analysed by transmitting signal with different polarisations from the femtocell BS against the random orientation MS. Literature on the QHA has also been reviewed. The QHA is a compact multi element antenna and capable of producing a high quality circular polarised signal over wide angle. The techniques to miniaturise, widen the bandwidth and provide multiband capability have been reviewed. The beam steering capability is also important so that the femtocell can 2.6 Summary 28 adjust its coverage to minimise inter-cell interference and maximise data throughput. Two methods to enable the QHA to have beam steering capability have been identified. The first method is the optimum combining between the multiple elements of the QHA. Another promising method to provide low complexity beam steering capability is parasitic QHA. These methods have not been fully explored in the literature. Chapter 3 Evaluation of Base Station Antenna with Different Polarisations in a Small Cell Network The tremendous growth in mobile data traffic in the past ten years creates the need for a cellular network with higher spectral efficiency. One method to achieve this is by reducing the cell size to increase spatial reuse [2] such as in a small cell network (SCN). One example of deploying a SCN is to use a public femtocell. Public femtocell is a short range and low power base station (BS) that could be randomly deployed in public places such as railway stations and shopping malls. By deploying many BSs in smaller cell size, higher throughput per user is achieved as the number of users per femtocell is low. Furthermore, with the reduction in transmit distance, a more reliable and stronger signal link between BS and mobile station (MS) will also ensure throughput improvement [2]. However, the reduction in cell size could also lead to higher inter-cell interference. Nowadays, the handset or mobile station (MS) is no longer mainly used for making calls (Talk Mode), but also for watching video and surfing the Internet (Data Mode). Therefore the user will handle the MS differently in these two situations as shown in Figure 3.1. In Talk Mode, the MS is inclined at angle α from the vertical position and held by the side of the head. In Data Mode, the MS is inclined at angle β from the horizontal position and held in front of the body. The inclination angle α and β varies from user to user [36]. Depending on the MS antenna design and usage orientation, the dominant polarisation could change in Talk and Data Modes causing polarisation mismatch. In Talk Mode, as the MS orientation is slanted, vertical and horizontal components exists. In Data Mode, the dominant polarisation might change to the orthogonal polarisation 30 (a) Talk mode (b) Data mode Figure 3.1 Mobile station orientations (Reproduce in this section for reference). as compared to the upright orientation. Under daily usage, antenna misalignment between the BS and MS due to human handling in these two modes could not be avoided. As BS antennas and MS antennas in the upright orientation are typically designed to be vertically polarised [157, 158], there will be a polarisation mismatch loss of up to 10-20 dB when the two antennas are not perfectly aligned [49], which could be possible for MSs in Data Mode. Signal from one polarisation will partially get depolarised into the orthogonal polarisation after multiple reflections on an oblique surface. This is typically the situation in a microcell environment, which helps in reducing polarisation mismatch caused by the MS antenna orientation. However, in a short range open space communication link, it is expected that the line-of-sight (LOS) between the BS and MS is usually available. Even when there is obstruction between the BS and MS, a dominant reflected signal will still reach the MS. Under these situations, a signal will still largely maintain its original polarisation state due to limited number of reflections. In this chapter, we investigate the performance of several MS receive antennas against the BS transmit antenna with different polarisation is investigated. Section 3.2 covers the evaluation in anechoic chamber as well as the effect of human body on the antenna performance. A field measurement campaign was performed and the result is discussed in Section 3.4. The evaluations in this chapter were performed at 2.4 GHz ISM band primarily due to two reasons. Firstly, this band is close to LTE 2600 MHz, LTE 2100 MHz and higher UMTS bands. The measurement results at 2.4 GHz will be closely relevant for these bands. Secondly, 2.4 GHz ISM band is license free and measurement permit was not required as long as the transmit power is lower than the allowed amount. This help in reducing the 31 3.1 Mobile stations’ Antennas (a) MS1 with meandered wire antenna (b) MS2 with PIFA Figure 3.2 Mobile stations. planning time and outdoor measurement can be performed whenever is needed . 3.1 Mobile stations’ Antennas Two commonly used antennas for the mobile stations are meandered wire antenna and planar inverted-F antenna (PIFA). These antennas are chosen in this work as they have orthogonal polarisations with each other in the upright orientation as oriented in Figure 3.1. The first mobile station, MS1 is equipped with a meandered wire antenna as shown in Figure 3.2(a). It is horizontally polarised in the upright orientation as can be seen from the radiation pattern plots in Figure 3.3. On the other hand, the second MS, MS2 as shown in Figure 3.2(b) has a vertically polarised planar inverted-F antenna (PIFA) in the upright orientation. Its radiation pattern is shown in Figure 3.4. These mobile stations are positioned in Talk and Data Modes, as operated by users. Both horizontal and vertical polarisation with respect to the base station exist when MSs are positioned at slanted angle in Talk Mode. On the other hand, in Data Mode, the primary polarisation for MS1 and MS2 are horizontal and vertical, respectively. The effect of human body on antenna performance is also evaluated such as holding the antenna in hand, in front of the body and by the head. 32 3.1 Mobile stations’ Antennas 0o 0o 10 dB −30o 10 dB 30o −30o 0 −60o 0 60o −10 −60o −20 −90o 90o −90o 90o −20 −20 −10 −10 −120o 120o −120o Vertical pol. Horizontal pol. 0 −150 60o −10 −20 o 30o 10 dB 120o Vertical pol. Horizontal pol. o o 150 −150 o 0 10 dB 150o o 180 180 (a) Azimuth (θ = 90°) (b) Elevation (φ = 0°) Figure 3.3 Radiation pattern of the MS1 . 0o 0o 10 dB −30o 10 dB 30o −30o 0 −60o −60o 60o −10 60o −10 −20 −20 −90o 90o −90o 90o −20 −20 −10 −10 −120o 120o Vertical pol. Horizontal pol. 0 −150o 30o 0 10 dB 150o −120o 120o Vertical pol. Horizontal pol. −150o 0 10 dB 150o 180o 180o (a) Azimuth (θ = 90°) (b) Elevation (φ = 0°) Figure 3.4 Radiation pattern of the MS2 . 33 0 0 −5 −5 −10 −10 Magnitude, dB Magnitude, dB 3.2 The Effect of Human Body on the Mobile Stations’ Antennas Performance −15 −20 Free space Phantom hand Human hand Phantom hand and head Human hand and head −25 −30 2 2.2 2.4 2.6 Frequency, GHz 2.8 (a) MS1 (Meandered wire antenna) −15 −20 Free space Phantom hand Human hand Phantom hand and head Human hand and head −25 3 −30 2 2.2 2.4 2.6 Frequency, GHz 2.8 3 (b) MS2 (PIFA) Figure 3.5 Mobile stations’ return loss, S11 in various environments. 3.2 The Effect of Human Body on the Mobile Stations’ Antennas Performance In order to evaluate the impact of human tissue on the performance of the mobile stations, measurements of the return loss (scatttering parameter, S11 ) were performed when the mobile stations were held in the hand only and when the mobile stations were held by hand beside the head. Apart from the measurements with human hand and head, the measurements were also performed with a phantom hand and head. The phantom hand was constructed from latex glove which was filled with 1 % Sodium Chloride (NaCl) solution so as to mimic the conductivity of human hand. The conductivity of human hand, σh at 2.45 GHz was approximately 1.64 S/m [159]. The NaCl solution was made by dissolving 10 g of NaCl into 1000 mL of distilled water [160]. The phantom head was filled with glucose solution to mimic the permittivity and conductivity of the human head as described in [155]. The return loss measurements of mobile stations were also performed as references. Figure 3.5(a) shows the return loss measurements for MS1 which was equipped with meandered wire antenna. In free-space, MS1 resonated at 2.44 GHz with bandwidth of 160 MHz. When held with human hand, the resonant frequency increased by 20 MHz without significantly impacting the bandwidth. In human hand-head configuration, the resonant frequency decreased by 40 MHz. The return loss was about 20 dB lower at the resonant frequency and the bandwidth improved to 220 MHz. When holding by hand only, the ground plane at the bottom portion of MS1 was masked by the hand while the antenna at the top of MS1 was further away from the hand and as a result, there was minimal interaction between 3.2 The Effect of Human Body on the Mobile Stations’ Antennas Performance 34 the hand and the antenna. However, when the MS1 was held in hand-head configuration, a larger proportion of the ground plane was masked and at the same time, the head was closer to antenna. The head also has higher permittivity and conductivity compared to the hand. Apparently, the complex interaction between the antenna and human body changed the antenna’s input impedance, which in this case improved the return loss and bandwidth. Figure 3.5(b) shows the return loss measurements for MS2 which was equipped with PIFA. MS2 has a resonant frequency of 2.45 GHz with bandwidth of 130 MHz in free space. Holding the MS in human hand shifted the resonant frequency down by 50 MHz without significant impact on the bandwidth. However the return loss increased by 20 dB at the resonant frequency indicating input impedance change. In human hand-head configuration, the resonant frequency shifted up by 20 MHz. The return loss increased by about 23 dB at the resonant frequency which in turn limits the bandwidth to 40 MHz only. As the antenna of MS2 was positioned in the middle of the casing, the whole antenna was masked by the hand. The antenna’s near field was within a few millimetres from the hand, which will severely impact the performance of the antenna. The impact was more pronounced when the MS2 was held in hand-head configuration. From the return loss measurements of MS2 , it can also be seen that the measurements conducted using a phantom hand and hand-head configurations agree well with their human counterpart which is represented by the solid purple line in Figure 3.5(b). This indicates that the phantom models were closely resemble the human head and hand. However, measurements of the return loss for MS1 have shown inconsistencies between the phantom and human measurements. The inconsistencies could be attributed to the inconsistent placement of hand and head in those measurements. Even the location of fingers on the mobile stations could affect the measurements. In MS2 measurements, the hand was masking the whole antenna as discussed in the previous paragraph. As a result, the antenna’s performance was less sensitive to the hand and head placements as compared to MS1 measurements where the hand is holding the ground plane area only. Studies by other researchers on the effect of human body in proximity to MS can be found in [161–165]. The list is not comprehensive but they serve as a good comparison to the result in this section. The results from those reports has shown that, in proximity with human tissue, there will be change in return loss and as resonant frequency. In general, the result in this section and from those reports agree well. The main factors that determine the severity of the performance impact are distance from of the human tissue the antenna, antenna design and placement of the antenna on MS. 3.3 Performance Measurement of BS Transmit Antenna with Different Polarisations in Anechoic Chamber. 35 3.3 Performance Measurement of BS Transmit Antenna with Different Polarisations in Anechoic Chamber. In this subsection, we investigated the performance of MS1 and MS2 orientations against transmit antennas at the base station (BS) with different polarisations in anechoic chamber. Anechoic chamber is a type of an indoor test range. The measurements were conducted in the anechoic chamber where there is very low reflection from the wall and floor. The anechoic chamber also prevents any interference from outside the chamber from contaminating the measurement. In order to evaluate the effect of BS antenna polarisation, a linear polarised (LP) horn antenna and a circular polarised (CP) spiral conical antenna were used as the transmitters. The horn antenna was oriented to achieve vertical polarisation (VP), horizontal plarisation (HP) and 45° slanted (SP) polarisations. For the measurements using the LP and CP antennas, the measurement equipments were calibrated separately as the two types of antennas have different gain. The performance of the four BS transmit antenna polarisations were measured against MS1 and MS2 in Talk Mode as well as in Data Mode. The positioning of the mobile stations in Talk Mode and Data Modes are as shown in Figure 3.1(a) and Figure 3.1(b), respectively. The inclination angles, α and β were fixed to about 60° and 25°, respectively. The radiation pattern measurements were performed with and without human interaction i.e. holding the mobile stations. With the human body positioned along θ = 0° axis and facing φ = 0°, the radiation pattern measurements were performed on the φ -plane at θ = 90°. In Talk Mode, the mobile stations were held by the hand at the left side of the head (φ = 270°). The radiation pattern measurements are summarised in Figure 3.6 and Figure 3.7 for MS1 and MS2 , respectively. The left and right sub-figures show the radiation pattern measurement without and with human interactions, respectively. Talk and Data Modes are shown in the top and bottom sub-figures, respectively. In reference to Figure 3.6, without human interaction, MS1 has generally higher gain for vertical polarisation compared to horizontal polarisation in Talk Mode. However, in Data Mode, this situation is reversed where HP gain is higher than VP gain. The gain in HP is 11 dB and 4 dB higher than VP at φ = 0° and φ = 270°, respectively for MS2 in Talk Mode. However, in Data Mode, gain in VP is higher than HP by 5 dB and 9 dB at φ = 0° and φ = 180°, respectively. The gain for SP and CP is close to the gain of the dominant polarisation. With human interaction i.e. holding the mobile stations in Talk or Data Modes, part of the radiated signal is absorbed and reflected by the human body. In Talk Mode, the radiation 3.3 Performance Measurement of BS Transmit Antenna with Different Polarisations in Anechoic Chamber. 36 0o 0o 5 dB o −30 5 dB o 0 o 30 −30 −5 o −60 −10 o 60 −15 −20 −20 −25 −25 −90o 90o −90o 90o −25 −25 −20 −20 −15 −15 −10 120o −5 VP HP SP CP 0 −150o 5 dB 60o −60 −15 −120o 30o −5 −10 o 0 150o −120o 120o −10 −5 0 −150o 5 dB 150o 180o 180o (a) Talk Mode (without human interaction) (b) Talk Mode (with human interaction) 0o 0o 5 dB −30o 0 5 dB 30o −30o −5 60o −10 −60o −15 −20 −20 −25 −25 90o −90o 90o −25 −25 −20 −20 −15 −15 120o −10 −120o −5 0 5 dB 120o −10 −5 −150o 60o −15 −90o −120o 30o −5 −10 −60o 0 0 150o o −150o 5 dB 150o o 180 180 (c) Data Mode(without human interaction) (d) Data Mode (with human interaction) Figure 3.6 MS1 radiation pattern without and with human interaction on the φ plane at θ = 90° in Talk (α = 60°) and Data (β = 25°) Modes. 3.3 Performance Measurement of BS Transmit Antenna with Different Polarisations in Anechoic Chamber. 37 0o 0o 5 dB o −30 10 dB o 0 o 30 0 −5 −10 o 30o −30 60o −60 −60o 60o −10 −15 −20 −20 −25 −90o 90o −90o 90o −25 −20 −20 −15 −10 −120o −10 120o −5 VP HP SP CP 0 −150o 5 dB 150o −120o 120o 0 −150o 10 dB 150o 180o 180o (a) Talk Mode (without human interaction) (b) Talk Mode (with human interaction) 0o 0o 10 dB −30o 5 dB 30o −30o 0 −60o 0 30o −5 60o −10 −10 −60o 60o −15 −20 −20 −25 −90o 90o −90o 90o −25 −20 −20 −15 −10 −120o 120o −120o 120o −10 −5 0 0 o −150 10 dB 150o o −150o 5 dB 150o o 180 180 (c) Data Mode(without human interaction) (d) Data Mode (with human interaction) Figure 3.7 MS2 radiation pattern without and with human interaction on the φ plane at θ = 90° in Talk (α = 60°) and Data (β = 25°) Modes. 3.3 Performance Measurement of BS Transmit Antenna with Different Polarisations in Anechoic Chamber. 38 towards the right side of the head (φ = 90°) is attenuated due to absorption and reflection by the head. Attenuation of as high as 30 dB could be seen at φ = 90° for both MS1 and MS2 . In Data Mode, the radiation towards the back of the body (φ = 180°) is attenuated by as high as 25 dB due to absorption and reflection of the human body. In Talk Mode, the radiation is reflected towards the left side of the head. There is a gain improvement as compared to when the mobile stations were in free-space as the overall radiation is towards the left side of the head. The maximum gain occurs at approximately φ = 240° for both MS1 and MS2 in the presence of the head in Talk Mode. With human interaction, there is gain improvement at the direction of maximum gain (φ = 240°) for most polarisations, as can be seen in Figure 3.8(a) and Figure 3.8(b) for MS1 and MS2 , respectively. For example, there is about 7 dB gain improvement with human interaction as compared to the case without human interaction for MS1 with horizontal polarisation (HP) in Talk Mode. In Data Mode, the gain improvement for most polarisations in MS1 is not as significant except for VP, as can be seen in Figure 3.8(c). There is gain reduction of about 4 dB for each polarisation with human interaction holding the antenna in front of the body as can be seen in Figure 3.8(d) for MS2 . The gain reduction is mainly caused by the higher return loss when the mobile station is held by the human hand. The complex interaction between the radiating and reflected radiation could also attenuate the signal. In reference to Figure 3.8, the interaction with human body can also change the dominant polarisation of the mobile station. Without human interaction in Talk Mode, the dominant polarisation for MS1 is VP but it changes to HP with human interaction. Under human interaction, the gain for MS1 with HP is 3.4 dB higher than VP. The dominant polarisation for MS2 , changes from HP without human interaction to SP with human interaction. HP is 4.2 dB better than VP for MS2 in Talk Mode with human interaction. In Data Mode, the change of dominant polarisation is not observed where the dominant polarisation for MS1 and MS2 are HP and VP, respectively. Under human interaction for MS1 , the HP gain is 9.7 dB higher than VP while for MS2 , the VP gain is 4 dB higher than HP. For MS1 in Talk and Data Modes, the performance of SP and CP are about the same and their gain are between the HP and VP gain. In Talk Mode for MS2 , SP is the best and CP is the worst among the four polarisations tested. However, in Data Mode for MS2 , SP performance is in between HP and VP while CP is the worst among them. The difference in radiation mechanisms between SP and CP could lead to different levels of polarisation mismatch between the BS and MSs. In SP, the power is fully concentrated along the slanted axis while in CP the power is uniformly divided into two orthogonal axes. Without multipath propagation to depolarise the transmitted signal in free space, SP radiation could possibly 0 10 −5 5 Gain, dBi Gain, dBi 3.3 Performance Measurement of BS Transmit Antenna with Different Polarisations in Anechoic Chamber. 39 −10 −15 0 −5 Free space With human −20 VP HP SP BS polarisation Free space With human −10 CP (a) MS1 in Talk Mode (φ = 240°) VP HP SP BS polarisation CP (b) MS2 in Talk Mode (φ = 240°) 5 0 0 −5 Gain, dBi Gain, dBi −5 −10 −10 −15 −15 −20 −25 Free space With human VP HP SP BS polarisation (c) MS1 in Data Mode (φ = 0°) CP Free space With human −20 VP HP SP BS polarisation CP (d) MS2 in Data Mode (φ = 0°)) Figure 3.8 Measured MS antennas gain in the respective modes and directions without (in free space) and with human interaction. 3.4 Field Measurement Campaign 40 yield better gain compared to CP in free space and static environment. 3.3.1 Summary An analysis of the return loss measurement has shown that interaction with the human body could affect the performance of the MSs’ antennas in terms of resonant frequency, return loss level and bandwidth. The level of performance impact is determined by the level of human interaction and design of the antenna. Holding the mobile station in the hand has less impact on the antenna’s performance compared to holding in hand against the head. The design and placement of the antenna on the MS will also impact the antenna performance as it will determine the distance to the hand and thus, the masking effect of the hand. This is especially important when designing mobile station with multiple antennas as space is limited. An analysis on the performance of BS antennas with different polarisations against the mobile stations in anechoic has shown that neither vertically nor horizontally polarised BS transmit antennas are optimum to provide the best performance in Talk Mode and Data Mode for the two different antenna designs. For MS1 , which has the meandered wire antenna, the dominant polarisation in both Talk and Data Mode when held by human is horizontal polarisation. On the other hand for MS2 , the dominant polarisation when held by human is vertical and slanted polarisation in Talk Mode and Data Mode, respectively. There could be up to 10 dB gain difference between VP and HP. Since the result here is based on measurement in free space and under static condition, these situations could change once the depolarisation effect takes place in real environment, which is investigated during the field measurement campaign discussed in the next section. 3.4 3.4.1 Field Measurement Campaign Measurement site Measurements were carried out in the open concourse area of the School of Management building at the University of Surrey. Its environment resembles short range public femtocell deployment in large open space. It has a length of 38 m with a width of 6 m. Its height extends into the second floor of the building. Bridges on the first and second floor at the centre of the concourse area separates it into two sides. Measurements were carried out on the ground floor at the north-western side of the concourse area which is shown as the 41 3.4 Field Measurement Campaign (a) Floor plan [166] (b) Concourse area Figure 3.9 Measurement site at the School of Management building, University of Surrey. shaded area of Figure 3.9(a). A photograph of the concourse area is shown in Figure 3.9(b) where the red lines shows the distance covered by the subject during measurements. 3.4.2 Measurement Setup Four transmitters were used at the BS consisting of vertically polarised (VP), horizontally polarised (HP), circularly polarised (CP), and 45° slanted polarised (SP) antennas. Square patch antennas, oriented accordingly were used for the VP, HP, and SP antennas. The photograph and radiation pattern for the square patch antenna is shown in Figure 3.10. A quadrifilar helix antenna (QHA) [128] is used for the CP antenna and its radiation pattern is shown in Figure 3.11. The transmitters were positioned on the bridge of the first floor as indicated in Figure 3.9(b). Its height above the ground floor is 5.8 m. In order to ensure the measurement range is within the half power beam width (HPBW) of the antennas, the transmit antennas are down-tilted by about 20°. Two mobile stations or handsets with different antennas were used as the mobile receivers. They were the same mobile station as used in the free space analysis in Section 3.1. The first MS, MS1 was equipped with a meandered wire antenna. It was horizontally polarised in the upright orientation as can be seen in Figure 3.3. The second MS, MS2 has a vertically polarised printed inverted-F antenna (PIFA) in the upright orientation. Its radiation pattern is shown in Figure 3.4. The height of the mobile stations above the ground floor depends on the usage mode as well as height of the user. It is expected that the mobile stations will be approximately 1.7 m and 1.2 m above the ground floor in Talk Mode and Data Mode, respectively. 42 3.4 Field Measurement Campaign (a) Photo 0o 0o 10 dB −30o 10 dB 30o −30o 30o 0 −60o 0 60o −10 −60o −20 −20 −90o −90o 90o 90o −20 −20 −10 −10 −120o 120o −120o 120o Vertical pol. 0 Horizontal pol. −150o 10 dB 60o −10 Vertical pol. 0 Horizontal pol. 150o −150o 10 dB o 150o o 180 180 (b) Elevation radiation pattern at φ = 0° (c) Elevation radiation pattern at φ = 90° Figure 3.10 Square patch antenna photo and radiation patterns. 0o 10 dB −30o 30o 0 −60o 60o −10 −20 −90o 90o −20 −10 −120o 120o Co−pol. 0 Cross−pol. −150o 10 dB 150o o 180 (a) Photograph (b) Elevation radiation pattern at φ = 90° Figure 3.11 QHA photograph and radiation pattern. 3.4 Field Measurement Campaign 43 Figure 3.12 Propsound MIMO Channel Sounder setup. An Elektrobit Propsound wideband MIMO channel sounder was used to capture the channel coefficient for all transmit and receive antenna combinations as shown in Figure 3.12. Measurements for all combinations are performed sequentially where the channel sounder utilised a fast switching technique through Time Division Multiplexing (TDM). The acquisition period for the channel matrix is much less than the coherent time of the channel and as such the channel could be considered constant. The channel from multiple BS antennas to the same MS could be measured and simultaneously compared using this technique. Measurements were performed at a centre frequency of 2.47 GHz with a bandwidth of 200 MHz, which is representative of Long Term Evolution (LTE) bands. The maximum walking speed during measurements was about 1 m s−1 . The maximum Doppler shift for this measurement was 8.23 Hz. According to Nyquist sampling criteria, the sampling frequency needs to be at least twice the maximum Doppler shift so that the signal will be accurately sampled. In this measurement, a sampling frequency of 32 Hz was used which exceed the Nyquist criteria. The narrowband channel data was extracted from the wideband data, which is recorded at distances from 6 m to 16 m from the BS. Both forward and backward facing measurements with respect to the BS scenarios were evaluated in Data Mode, thus effects of body loss were also accounted for. In Talk Mode, only forward facing measurements with respect to the BS scenario were evaluated. For the forward facing direction in Data and Talk Modes, five measurements were recorded for each mode and mobile station. For the backward facing direction in Data Mode, only one measurement was recorded. The measurement scenarios are summarised in Table 3.1. The positioning of the mobile stations in Talk Mode and Data Modes are shown in Figure 3.1(a) and Figure 3.1(b), respectively. In Talk Mode, the mobile phone is inclined at angle α of 44 3.4 Field Measurement Campaign Table 3.1 Measurement Scenarios and the number of measurement performed. Mode Data Mode (Forward facing) Talk Mode Data Mode (Backward facing) MS1 MS2 Obstruction 5 5 1 5 5 1 LOS OLOS OLOS approximately 60° from the vertical position and held by the side of the head. In Data Mode, the mobile station was inclined at angle β of approximately 25° from the horizontal position and held in front of the body. Interference from Wireless Local Area Network (WLAN) A back-to-back measurement from the transmitter to the receiver units was performed in the initial setup. This would compensate any loss or phase delay due to the equipment. In post processing, further compensation was need to take into account the transmit and receive antenna gain and cable loss. The measurement campaign were performed at the 2.4 GHz ISM band. There were instances where the captured signal is corrupted by the interference from WLAN signal. This could be easily observed in the Power Delay Profile (PDP) of the channel. When comparing the PDP of the channel without and with interference, it could be seen that the noise floor rise significantly. Based on the data without interference, the noise floor level was established and the time instances when the interference occurred were recorded. The narrrowband data was then processed to remove the interference based on the recorded time instances. When interference occurred, the narrowband channel samples were linearly interpolated from the clean samples before and after the samples affected by interference. In all measurements, the longest sample sequence with interference was less than five and the total number of interpolated samples in each measurement is less than 5 %. It is expected that interpolated samples will not significantly affect the distribution of the samples. 45 3.4 Field Measurement Campaign Channel Matrix The measured narrowband channel matrix, Hpolarisation for the four BS transmit antennas and two MS’s antennas is given by ' # hMS1 ,V P hMS1 ,HP hMS1 ,CP hMS1 ,SP Hpolarisation = hMS2 ,V P hMS2 ,HP hMS2 ,CP hMS2 ,SP (3.1) where the first and second subscript set of each channel coefficient, hrx,tx refer to the receive and transmit antennas, respectively. The analysis is only concerned with the performance of each transmit-receiver pair, consequently each channel coefficient, hrx,tx will be analysed separately. Due to the difference in the gain between the linear and circular polarised BS antennas, the magnitude of each channel coefficient, |hrx,tx |2 is normalised. The received power, Prx at the MS is given by Eq. (3.2). As the measurements were performed at a fixed transmit power, Ptx , it is sufficient to do the analysis based on the magnitude of channel coefficient, |hrx,tx |2 only. Prx = |hrx,tx |2 Ptx 3.4.3 (3.2) Result: Overall Characteristic Figure 3.13(a) shows a measurement sample of the magnitude of the channel coefficient, |hrx,tx |2 for MS1 , as transmitted by the four BS antennas in Data Mode. The data is then transformed into complementary cumulative distribution function (CCDF) plot, as shown in Figure 3.13(b). The plots correspond to the four transmitters at the BS; namely VP, HP, CP, and SP. In the example, shown in Figure 3.13(b), it can be seen that the HP antenna has the highest gain while the VP has the lowest gain. The SP is only slightly better than the VP and CP is slightly worse than HP. The CP signal is equally divided into two orthogonal axes while SP signal is concentrated on one slanted axis. It is evident that CP is better able to reduce polarisation mismatch loss between the BS and a randomly oriented MS. The magnitude of the channel matrix for VP base station antenna is taken as a reference in performance gain calculation. In the literature, the gain is typically reported at the 95 % or 99 % probability levels. However, due to a limited number of samples at low signal levels, the gain at these probability levels is not representative of the overall gain in certain 46 3.4 Field Measurement Campaign −40 −60 −70 −80 VP HP CP SP −90 −100 Data points (a) Plot of |hMS1 ,tx |2 over time 0 50 % prob. |hrx,tx|2 > Abscissa 1 |hMS ,tx|2, dB −50 80 90 99 VP HP CP SP 99.9 −100 −90 −80 |h −70 −60 |2, dB −50 −40 rx,tx (b) CCDF plot Figure 3.13 Sample measurement data (MS1 , Data Mode). 47 3.4 Field Measurement Campaign 12 6 HP CP SP 8 6 4 2 0 4 Relative gain over VP, dB Relative gain over VP, dB 10 HP CP SP 2 0 −2 −4 User 1 User 2 User 3 User 4 User 5 (a) MS1 (Meandered wire antenna) Average −6 User 1 User 2 User 3 User 4 User 5 Average (b) MS2 (PIFA) Figure 3.14 Relative gain variations due to users in Talk Mode. cases. In this work, the gain achieved using other polarisations is reported as the gain at 80 % probability level over the reference VP antenna. The comparisons in Figure 3.13(b) show the improvement or deterioration of the channel by using different polarisations over a VP base station antenna. Variation due to users Figure 3.15 and Figure 3.14 show the relative gain of using different polarisations transmit antennas over a VP base station antenna for all forward facing measurements to BS in Talk Mode and Data Mode, respectively. It can be seen from the measurement results that, the gain varies from user to user. This is mainly due to the differences in the way the mobile stations were handled by the users. The inclination angle α or β and also the height from floor varied among different human subjects. Hand grip variation when holding the mobile stations could also cause the gain variations [36]. In Figure 3.14, it can be seen that all measurements for MS1 a produced positive gain of HP over VP base station antenna in Talk Mode. In contrast to MS1 , there can be signal gain or loss of HP over VP in MS2 . This can be attributed to the inclination angle, α (Figure 3.1a) and the cross polarisation level of the mobile station antennas. The inclination angle varies among users but it has a typical value of 60° [36]. Cross polarisation level in MS1 is higher than MS2 , at certain positions, as can be seen in Figure 3.3 and Figure 3.4. It is expected that the dominant polarisation of the MS antenna (with respect to the BS) will change depending on the inclination angle. MS2 is vertically polarised in the upright orientation and the change of the dominant polarisation (with respect to the BS) is evident 48 3.4 Field Measurement Campaign 12 6 HP CP SP 8 6 4 2 0 4 Relative gain over VP, dB Relative gain over VP, dB 10 HP CP SP 2 0 −2 −4 User 1 User 2 User 3 User 4 User 5 Average −6 User 1 (a) MS1 (Meandered wire antenna) User 2 User 3 User 4 User 5 Average (b) MS2 (PIFA) Figure 3.15 Relative gain variations due to users in Data Mode (Forward facing). in Figure 3.14(b) (MS2 in Talk Mode). Positive gain of HP over VP indicates that MS2 , as held by the three users was dominantly horizontally polarised (with respect to the BS). The other two users have negative HP gain over VP indicating that MS2 was dominantly vertically polarised. However, this behaviour was not apparent for MS1 in Talk Mode where all measurements show positive gain for HP signal indicating that MS1 is dominantly horizontally polarised in that orientation. This is due to the higher level of cross polarisation in the MS1 meander wire antenna. Similar trends can be observed for Data Mode as shown in Figure 3.15. Figure 3.16 summarises the gain over VP BS antenna for all scenarios. For forward facing Data and Talk Modes, the average gain is reported. Even though only one measurement was carried out for each MS in backward facing Data Mode, the result is included as well as a comparison. However, care must be taken in interpreting this particular result as it might not give a good overall representation of that mode. Forward Facing Data Mode/ Line-of-sight(LOS) It can be seen from the forward facing Data Mode chart in Figure 3.16(a) that, the average gain increase is 8.6 dB, 6.3 dB and 4.6 dB for HP, CP and SP, respectively over VP as base station transmit antenna for MS1 . The average gain over VP for MS2 , is −2.8 dB, −0.2 dB and −0.3 dB for HP, CP and SP, respectively. Negative gain indicates a loss or weaker signal. The large gain of HP over VP in MS1 is due to the availability of a LOS link between the BS and the MS, where level of signal depolarisation is low. Even though the LOS link 49 3.4 Field Measurement Campaign 12 6 HP CP SP 8.6 8 6.3 6 4.6 3.9 4 3.9 3.2 3.7 2.7 3.0 2 0 4 Relative gain over VP, dB Relative gain over VP, dB 10 HP CP SP 2.4 2.7 2 0 −0.2 −0.3 Data Mode (Forward) 3.1 1.9 2.3 0.4 Data Mode (Backward) Talk Mode −2 −2.8 −4 Data Mode (Forward) Data Mode (Backward) −6 Talk Mode (a) MS1 (Meandered wire antenna) (b) MS2 (PIFA) Figure 3.16 Summary of relative gain over vertically polarised (VP) BS antenna. also exists in the MS2 case, the boresight of the MS2 is pointing toward the operator causing smaller gain between HP and VP base station antennas. While the HP base station antenna is favourable for MS1 , the VP base station antenna is favourable in MS2 , for the forward facing Data Mode. These can be attributed to the dominant polarisation state of the mobile stations antennas (with respect to BS) in Data Mode orientation. From Figure 3.3 and Figure 3.4, it can be seen that the MS1 is horizontally polarised and MS2 is vertically polarised in the Data Mode orientation. The performance of CP and SP are in between HP and VP in MS1 as well as MS2 . In MS1 , CP has a gain of 6.3 dB, −2.3 dB and 1.7 dB over VP, HP and SP, respectively. In MS2 , the performance of CP is similar to that of VP and SP while being better than HP by 2.6 dB. Backward Facing Data Mode and Talk Mode/ Obstructed-LOS (OLOS) Backward facing Data Mode and forward facing Talk Mode represent OLOS scenarios. The human body is blocking the LOS link between MS and BS in backward Data Mode. In forward facing Talk Mode, the head is partially obstructing the LOS link between the MS and BS. These obstructions are causes the signal to take an indirect route before reaching the mobile stations. In this case, signal depolarisation occurs due to reflections from the surrounding scattering and human body. The effect of signal depolarisation is observed when comparing the result of the OLOS to LOS scenarios, in Figure 3.16. In MS1 (Figure 3.16(a)), the gain of HP over VP is 8.6 dB in forward Data Mode. However, in backward facing Data Mode and forward facing Talk Mode, the gain is reduced to 3.9 dB and 2.7 dB, respectively. 50 3.4 Field Measurement Campaign −50 Signal Local Mean Path Loss −55 |hrx,tx|2, dB −60 −65 −70 −75 −80 8 9 10 11 12 13 Distance, m 14 15 16 17 18 Figure 3.17 Radio signal components of a sample measurement data. A similar result could be observed for MS2 , in Figure 3.16(b). In forward facing Data Mode, the absolute difference between is HP and VP is 2.8 dB and in the backward direction it is only 0.4 dB. In the Talk Mode, the gain of HP over VP is 2.8 dB. The reason for the positive gain has been explained previously. The gain of CP over VP in MS1 and MS2 are 3.7 dB and 3.1 dB, respectively. In comparison to HP and SP, CP is about 1dB better in both mobile stations. This data is also important in deriving the empirical path loss model for public femtocell. 3.4.4 Result: Small Scale Fading (Fast Fading) The small scale characteristic or fast fading (solid line) is the rapid fluctuation of the radio signal around the local mean (dash-dot line), as shown in Figure 3.17. The local mean, m(x) is found by averaging the radio signal over a short distance as given by 1 x+L m(x) = ∑ s(x) 2L x−L (3.3) where L is the length samples where local mean is averaged and s(x) is the radio signal [167]. The fast fading component, r(x) is extracted from the overall radio signal by using Eq. (3.4) [167]. r(x) = s(x) m(x) (3.4) When a direct or line-of-sight (LOS) component is available, the distribution of the fast 51 3.4 Field Measurement Campaign fading signal can be approximated by a Ricean distribution. The probability density function (p.d.f.) of a Ricean distribution is given by f (r|a) = r − r2 +a22 ra e 2σ I0 ( 2 ) 2 σ σ (3.5) where r is the magnitude of the complex envelope, a is magnitude of the LOS signal, 2σ 2 is the average power of the multipath component and I0 is the modified Bessel function of first kind and zero order. If the LOS signal diminishes and only the multipath signal is available, the distribution of the fast fading signal will approach the Rayleigh distribution with the p.d.f. given by: f (r|a) = r − r22 e 2σ σ2 Krice = 10 log10 a2 2σ 2 (3.6) (3.7) The Rice K-factor, as defined by Eq. (3.7), is commonly used to describe the fast fading signal. It is the ratio of the power of the LOS signal or coherent component, a2 to the power of the incoherent or multipath component, 2σ 2 . The Rice K-factor can be estimated by using distribution fit method such as Maximum Likelihood Estimate (MLE) and then Kolgomorov-Smirnov-test (KS-test) can be used for verification. A simpler estimation method as evaluated in [168] has been shown to provide good result. It is a moment-based estimator and the Rice K-factor can be found by using Eq. (3.8) where γ is given by Eq. (3.9), V [.] is the variance operator and E[.] is the mean operator. ′ Krice √ 1−γ √ = 10 log10 1− 1−γ (3.8) V [r(x)2 ] E[r(x)2 ]2 (3.9) γ= In this work, there are five measurements for each mode and mobile station in the forward facing direction and one measurement in Data Mode for the backward facing direction with respect to the base station. By using the moment-based estimator, described above, the 3.4 Field Measurement Campaign 52 Figure 3.18 Summary of Rice K-factor. Rice K-factor is calculated from the channel coefficient, as summarised in Figure 3.18. For the measurements in the forward facing direction, the average Rice K-factor is reported as well. By comparing the forward and backward facing measurements in Data Modes, it can be seen that there is a significant different in the Rice K-factor values. This is especially true for MS1 in Data Mode where on average, the difference is about 5 dB. The average Rice K-factor difference for MS2 is about 3 dB between the forward and backward facing directions. The lower Rice K-factor in the backward facing direction is due to the lower power of the coherent signal as the signal is blocked by the human body. Even though the Rice K-factor is lower in the backward facing direction, it still indicates that the coherent or LOS signal is still largely available. This validates the assumption that the surrounding scattering provides an indirect path for a dominant strong signal to reach the mobile stations from the base station even when it is blocked by the human body. The multipath echoes are weaker in comparison to the dominant signal from the indirect path. In the forward facing direction, the Rice K-factors are slightly higher in Data Mode as compared to Talk Mode. In the Data Mode, the mobile stations have an unobstructed link to the base station. However, in Talk Mode, the link to the base station is partially blocked by the head. From the mobile stations’ point of view, it can be seen that MS1 has higher Rice K-factors compared to MS2 in both Talk and Data Modes. This could be attributed to the location of the antennas on the mobile stations. The antenna’s location on MS1 at the top of the casing provides a better link to the base station as compared to MS2 where the antenna is placed at the middle of the casing. The difference of Rice K-factor among different polarisations is clearly observed. This is expected as the mobile stations has the same link to the BS transmit antennas. 53 3.4 Field Measurement Campaign 3.4.5 Large Scale (Slow Fading) Characteristic A radio signal can be decomposed into path loss, large scale fading (slow fading) and small scale fading (fast fading) components, as shown in Figure 3.17. The fast fading signal is the rapid fluctuation of the overall radio signal around the local mean. The local mean fluctuates as a function of distance. The fluctuation of the local mean is known as large scale or slow fading. Slow fading characteristic will depends on its propagation surrounding as shadowing characteristics will be different. Path loss (PL) is the signal decay due to the radial distance between the transmitter and receiver. It depends on the frequency and propagation medium. The path loss, PL and its relationship with frequency and propagation medium is governed by PLdB = 20 log10 fc + NPL log10 DPL + L f − 28 (3.10) where fc is the frequency (in MHz), NPL is distance power loss coefficient, DPL is separation distance (in metre) and L f is the floor penetration loss factor loss (in dB). The floor penetration loss factor is not applicable in these measurements as mesaurement is not performed through floor. By using the maximum likelihood estimate (MLE), the path loss is estimated from the local mean, based on the simplified path loss equation, Eq. (3.11). In the equation, aPL approximately accounts for the frequency dependent and the constant factors. ′ = aPL + NPL log10 DPL PLdB (3.11) SFdB = M(x)dB − PL′ (3.12) The slow fading component (in dB), SFdB is calculated by subtracting the estimated path loss, PL′ from the local mean in dB, M(x)dB , as shown in Eq. (3.12). The slow fading distribution is log-normal when expressed in linear term, and normal when expressed in Decibels. Based on the distribution of slow fading component in Decibels, the standard deviation is extracted and the result is summarised in Figure 3.19. There are five measurements for each mode and mobile station in the forward facing direction and one measurement in Data Mode for the backward facing direction with respect to the base station. For the measurements in the forward facing direction, the average Rice K-factor is reported. From the result in Figure 3.19, there is no tangible evidence of significant variation of the slow fading distribution among all cases. The standard deviation, among Data Modes and Talk Modes are about the same. There is also no significant standard deviations difference 54 3.5 Summary Figure 3.19 Summary of slow fading standard deviation, σSF . among the different polarisations. Standard deviation in all cases varies between 1.3 dB to 3.8 dB. The overall average standard deviation is about 2.6 dB. The variation of slow fading at this level is as expected for the public base station environment. 3.5 Summary Measurement in Anechoic Chamber An analysis of return loss measurement has shown that interaction with human body could affect the performance of the mobile stations’ antennas in term of resonant frequency, return loss level and bandwidth. The level of performance impact is determined by the level of human interaction and the design of the antenna. Holding the mobile station in the hand has less impact on the antenna’s performance compared to holding it in hand against the head. The design and placement of the antenna on the mobile stations will also impact the antenna performance as it will determine the distance to the hand and thus, the masking effect of the hand. This is especially important when designing mobile station with multiple antennas as space is limited. The analysis on the performance of BS antennas with different polarisations against the mobile stations in free space has shown that neither vertically nor horizontally polarised BS transmit antennas are optimum to provide the best performance in Talk Mode and Data Mode, for the two different antenna designs. For MS1 which has the meandered wire antenna, the dominant polarisation in both Talk and Data Mode when held by human is horizontal polarisation. However, the dominant polarisation when held by human for MS2 is vertical and slanted polarisation in Talk Mode and Data Mode, respectively. There could be 3.5 Summary 55 up to 10 dB gain difference between VP and HP. Field Measurement The effects of BS antenna polarisations in LOS and OLOS scenarios have been evaluated in a public femtocell environment. Evaluations were carried out in Data and Talk Mode for mobile stations with horizontally and vertically polarised antennas. The analysis of small scale and large scale fading has shown that these parameters are not dependent on the polarisation of the transmit signal. All polarisations performed similarly in this aspects. The analysis of the overall signal characteristics has shown that the polarisation of the mobile station with respect to the base station is random, depending on the users handling and the original polarisation of the mobile station. In such environments, VP or HP base station antennas will only be advantageous to certain users while at the same time they would deteriorate the performance of other users. Neither VP nor HP base station antenna are optimum to accommodate all scenarios. It is found that circularly polarised (CP) base station antenna is a good compromise and is more effective in reducing polarisation mismatch loss due to human handlings among other polarisations evaluated under most scenarios. In MS1 forward facing Data Mode, the gain of CP over VP, HP and SP are 6.3 dB, −2.3 dB and 1.7 dB, respectively. In MS1 Talk Mode, the gain of CP over VP, HP and SP are 3.7 dB, 1.0 dB and 0.7 dB, respectively. For MS2 in forward facing Data Mode, the gain of CP over VP, HP and SP are −0.2 dB, 2.6 dB and 0.1 dB, respectively. For MS2 in Talk Mode, the gain of CP over VP, HP and SP are 3.1 dB, 1.2 dB and 0.8 dB respectively. Chapter 4 Controlling the Radiation Pattern of Quadrifilar Helix Antenna The analysis the previous chapter has shown that the circular polarisation antenna can be a good compromise as a femtocell base station (BS) antenna. Based on the result, the quadrifilar helix antenna has been selected as the candidate for the femtocell BS antenna. The QHA typically has wide half power beamwidth (HPBW) in excess of 120°. For a single BS operation, this can be an advantage as a single antenna can cover a wide area. However, with multiple BSs deployed in small cell network, it is more desirable to have a relatively narrower HPBW so that each base station can cover a more specific area. One method to reduce the QHA HPBW is by increasing the number of QHA helical turn. However, there is a limit to this before the mutual coupling between the helical elements start to adversely affect the performance of the antenna and another technique is required. In this chapter, the use of parasitic element to improve the QHA gain in the boresight direction will be explored. By imroving the boresight gain of the QHA, more power will be concentrated towards the boresight direction and as a result, the HPBW will be reduced. Furthermore, if the gain of the BS is increased, the BS can transmit using lower power and yet achieve the same coverage distance. Another feature, that is desirable for a femtocell BS in a SCN, is the ability to steer or tilt the beam towards the desired direction. This can benefit in terms of better coverage and reduction in interference. For a SCN network, low complexity and low cost solutions are required due to the high number of BSs involved in a single network. In this chapter, the switched parasitic QHA will be evaluated. The parasitic elements at the sides of the QHA gives the QHA the ability to steer the beam. At the end of the chapter, the ability of an independent-fed QHA will be analysed. Even though this technique is more complex than 4.1 Reference Quadrifilar Helix Antenna 57 the switched parasitic QHA, it will be covered to provide a comparison. 4.1 Reference Quadrifilar Helix Antenna Introduction The analysis in this chapter is based on the printed right-hand screw direction QHA with element length of approximately 3/4-wavelength. With this winding configuration, the radiation of the QHA will be predominantly left-hand circular polarised (LHCP). The QHA is fabricated from R/flex 2005K828 flexible circuit material which is manufactured by Rogers Corporation. The substrate is a 50 µm-thick Kapton Polymide with relative permittivity, εr of 3.8 and loss tangent, tan δ of 0.01. Both sides of the substrate have 17.5 µm-thick copper backing. The detail specification of this material can be found in [169]. The QHA helical element design is printed onto one side of the flexible circuit material. The unnecessary copper section as well as the copper backing on the other side of the circuit material are etched away, during the fabrication process. The QHA in the unwrapped form is shown in Figure 4.1(a) where the copper section and substrate is shown in grey (darker shade) and pink (lighter shade), respectively. It can then be wrapped to form a cylindrical structure with radius, Rqha , as shown in Figure 4.1(b). Each helical element consists of a middle element, Lmiddle , a bottom stub, Sbottom and a top stub, Stop . The top and bottom stubs are not required for the functionality of the QHA but they will help in the design and fabrication processes. The total length of a helical element, Lelement is the sum of these three components, as given by Eq. (4.1). The slant or pitch angle, αqha of the middle helical section is given by Eq. (4.2) where N is number of turn in the middle section of the helical and Rqha is the radius of the QHA. The height of the QHA or axial length, Lax can be calculated from Eq. (4.3). Finally, the width of the helical element is Welement . The physical dimensions of the QHA are given in Table 4.1, will be used as a reference for the analysis in this chapter. The chosen operating frequency of the antenna is in the 2.4 GHz band. The number of turn is chosen to be relatively high so that the antenna will have higher gain and smaller half power beamwidth (HPBW). Lelement = Stop + Lmiddle + Sbottom (4.1) 58 4.1 Reference Quadrifilar Helix Antenna (a) Unwrapped form (b) Wrapped form Figure 4.1 Quadrifilar helix antenna (QHA) in the unwrapped and wrapped forms. Table 4.1 QHA physical parameters Parameter Dimension Middle element length, Lmiddle Element width, Welement Top stub, Stop Bottom stub, Sbottom Radius, Rqha Number of turn, Nqha αqha = cos−1 ( 83.5 mm 2 mm 0.5 mm 1 mm 10 mm 1 2πRqha Nqha ) Lmiddle q 2 Lax = Lmiddle + (2πRqha Nqha )2 + Stop + Sbottom (4.2) (4.3) Simulation Software The QHA is simulated using Transient Solver in Computer Simulation Technology (CST) software. It is a volume based numerical technique which is based on Finite Integration 4.1 Reference Quadrifilar Helix Antenna 59 Technique (FIT). FIT discretizes the integral form of the Maxwell’s equations [170]. Using this technique, the simulated structure are divided into small cells. The simulated structure is divided into small cells and time signal is excited at the input port. The Maxwell’s equations are applied at each cell and the individual solutions are summed up to form a larger set of equations. The final result is obtained by solving the larger set of equations. Depending on the structure, Transient Solver is a time consuming simulation. Scattering Parameters (S-parameters) and Radiation Pattern Even though the substrate of the flexible circuit material is thin, it can affect the resonant frequency of the helical element. In order to obtain a better agreement between the simulation and measurement results, the substrate is included in the simulation model. In the simulation, the QHA is positioned such that the boresight is directed along the z-axis, Port 1 and 3 are aligned on the x-axis and Port 2 and 4 are aligned on the y-axis as indicated in Figure 4.1(b). The QHA sits above a circular ground plane with radius of 1.5Rqha and its top surface is centred at the origin. For clarity, the ground plane is omitted from Figure 4.1(b). Figure 4.2(a) shows the S-parameters of the QHA when Port 1 is excited while other ports are terminated with matched impedance. With this feeding configuration, the resonant frequency is 2.51 GHz with an impedance bandwidth of 181 MHz, as represented by the S11 plot. Due to the close spacing between the helical elements, the radiation from the excited element couples to other elements as indicated by the S21 , S31 and S41 plots. The QHA is symmetrical and the same S-parameters will be obtained when exciting other elements. The input impedance of each QHA element is approximately 50 Ω. When the helical elements are excited simultaneously, the signal from one helical element will couples to other helical element which gives rise to mutual coupling. The effective impedance at one excitation port becomes the sum of contributions from other ports [171] and as a result, the resonant frequency could shift. Figure 4.2(b) shows the S-parameters when the helical elements are excited with equal magnitude and quadrature phase phase input where elements 1 through 4 have relative phase of 0°, 90°, 180° and −90°, respectively. The reason of the choice of the relative phase configuration will be apparent later. With this excitation, the resonant frequency shifts to 2.44 GHz with no significant change to the impedance bandwidth. The left-hand circular polarisation (LHCP) elevation gain and phase at φ = 0° of each helical element, are shown in Figure 4.3(a) and Figure 4.3(b). From the phase plot, it can be seen that there is inherently, 90° phase difference, between each adjacent element and, 180° phase difference, between non-adjacent elements along the z-axis (θ = 0°). When 60 0 0 −10 −10 Magnitude, dB Magnitude, dB 4.1 Reference Quadrifilar Helix Antenna −20 S11 −30 S21 S31 −40 −20 −30 S1 S2 S3 S4 −40 S41 −50 2 2.2 2.4 2.6 Frequency, GHz 2.8 3 −50 2 (a) Single element excitation 2.2 2.4 2.6 Frequency, GHz 2.8 3 (b) Quadrature excitation Figure 4.2 Scattering Parameters (S-parameters) the elements of the QHA are fed with equal phase, the radiation from one element will be canceled by the non-adjacent element. The radiation from each element could be coherently summed by feeding the QHA in equal magnitude and quadrature phase input where element 1 through 4 have relative phase of 0°, 90°, 180° and −90°, respectively. With this quadrature-phase feed, it can be seen, in Figure 4.4, that the phase of all elements are now equalised in the boresight direction. It can also be seen that the relative phase deteriorates at angles away from the boresight direction. The QHA radiation pattern and axial ratio is shown in Figure 4.5(a) and 4.5(b), respectively. The maximum circular polarised gain is 5.15 dBic with an axial ratio of less than 3dB between the elevation angle, θ of −77° and 77°. The half power beam width (HPBW) is 125°. Prototype of the Quadrature Feed Network(QFN) In the simulation, four input ports are used where each QHA element is fed separately with equal magnitude and 90°-phase difference. In order to simplify the feed network for the prototype antenna, a quadrature feed network (QFN) can be used. It has one input port and four output ports. A fabricated quadrature feed network based on dual-stage cascaded Wilkinson power divider and microstrip phase delay line is shown in Figure 4.6. The substrate is made from FR4 material with a thickness of 1.6 mm, relative permittivity, εr of 4.5 and loss tangent, tan δ of 0.0175. The backside of the feed network is backed by ground plane. From the input port, the first stage Wilkinson power divider uniformly divides the input signal into two. Each second stage Wilkinson power divider further divides the output of the 61 4.1 Reference Quadrifilar Helix Antenna 2 Element 1 Element 3 180 Element 2 Element 4 Relative phase, (in degree) 0 Gain, dBic −2 −4 −6 −8 −10 −180 −135 −90 −45 135 90 45 0 −45 −90 −135 Element 1 Element 3 −180 0 45 Theta, θo 90 135 180 −180 −135 −90 −45 (a) Gain Element 2 Element 4 0 45 Theta, θo 90 135 180 (b) Phase Figure 4.3 LHCP elevation radiation gain and phase for each helical element of the QHA. Relative phase, (in degree) 180 Element 1 Element 3 Element 2 Element 4 135 90 45 0 −45 −90 −135 −180 −180 −135 −90 −45 0 45 Theta, θo 90 135 180 Figure 4.4 LHCP elevation radiation phase of each element of the quadrature-fed QHA Ref. QHA (Co−pol.) o Ref. QHA (Cross−pol.) 0o10 dBic −30 5 o 30 0 4 60o −10 −20 −90o 90o −20 −10 −120o 120o Axial ratio, dB −60o 3 2 1 0 −150o 10 dBic 150o 180o (a) LHCP elevation radiation pattern 0 −90 −60 −30 0 30 Elevation angle , (θo) 60 (b) Axial ratio Figure 4.5 Elevation gain and axial ratio of the quadrature-fed QHA (φ = 0°). 90 4.1 Reference Quadrifilar Helix Antenna 62 Figure 4.6 Fabricated quadrature feed network. first stage Wilkinson power divider into another two uniform outputs. Microstrip lines with different length are used to provide the phase delay since each output has the same signal phase. Each microstrip phase delay line is configured such that each output has 90°-phase difference as required. In a similar way, this quadrature feed network can also be used to combine signal in receive mode. The simulated S-parameter for the QFN are shown in Figure 4.7. Ideally, when the input port (Port 5) of the QFN is excited, the output of the QFN at Port 1 to 4 should be −6 dB. However, due to losses and impedance mismatch in the microstrip circuit, the magnitude of the output is slightly lower as evidenced in Figure 4.7(a). Among the typical losses in microstrip circuit are dielectric, conductor and radiation losses. Not shown in the figure is the return loss of Port 5, which is about −28 dB at 2.45 GHz. It remains below −20 dB across the operating bandwidth of the reference QHA. It can also be seen in the figure that there is variation between the output magnitudes. This is largely due to the different configurations of each microstrip delay line. Each microstrip line could have different length and number of bend. The variation in the output magnitude could lead to higher axial ratio and cross-polarisation signal level. However, since the deviation is small, it is not expected to adversely affect the performance of the QHA. The phase of the transmission coefficient is shown in Figure 4.7(b) where the phase difference between the output ports are approximately 90°. Figure 4.7(c) shows the S-parameter when Port 1 is excited. The return loss (S11 ) of Port 1 is low and stays below −18 dB across the frequency range. The isolation between Port 1 and other output ports, as represented by S21 , S31 and S41 is very good at the centre frequency. When other output ports are excited, similar results, to those shown here, are 63 4.1 Reference Quadrifilar Helix Antenna 0 S15 150 S25 −2 S35 100 Relative phase Magnitude, dB S45 −4 −6 50 0 −50 S15 S25 −100 −8 S35 −150 −10 2 2.2 2.4 2.6 Frequency, GHz 2.8 3 2 S45 2.2 2.4 2.6 Frequency, GHz 2.8 3 (a) Magnitude of the S-parameter with excitation at (b) Phase of the S-parameter with excitation at Port 5 Port 5 0 Magnitude, dB −10 −20 −30 S11 −40 S 21 S31 S41 −50 S51 −60 2 2.2 2.4 2.6 Frequency, GHz 2.8 3 (c) Magnitude of the S-parameter with excitation at Port 1 Figure 4.7 Simulated S-parameter of the QFN. obtained. Prototype of the Quadrifilar Helix Antenna The QHA is fabricated from R/flex 2005K828 flexible circuit material, as discuss in the simulation setup section. From the flat form, the fabricated QHA is wrapped to form a cylindrical structure. For the purpose of validation, two types of feed network are used. They are an independent microstrip feed network (IMFN) and quadrature feed network (QFN), as can be seen in Figure 4.8. Each QHA element is soldered to the output ports of the feed network. The IMFN, shown in Figure 4.8(a), is fabricated from the same FR4 substrate material 64 4.1 Reference Quadrifilar Helix Antenna (a) QHA with independent ports (b) QHA with quadrature feed network Figure 4.8 Fabricated QHA with feed networks. as used for the QFN. It consists of four straight microstrip lines with 50 Ω input impedance. Each microtrip line is connected to one element of the QHA. The purpose of this feed network is to measure the S-parameter of one QHA element. When one port is excited, the remaining ports are terminated with matched impedance. The S-parameter measurement is performed using two-port vector network analyser (VNA). The measured S-parameter for Port 1 of the IMFN can be seen in Figure 4.9(a). From the return loss measurement (S11 ), the resonant frequency and bandwidth is found to be 2.52 GHz and 165 MHz. This is in a very close agreement with the simulation result. The deviation from the simulated result could be attributed to imperfections in the fabrication and construction processes as well as variations in the substrate material compared to the specification. The measured S21 , S31 and S41 are also in close agreement with the simulation result. Another aspect of the QHA that needs validation is the radiation pattern. The QHA is excited using the QFN, as shown in Figure 4.8(b). The measurement is performed in the anechoic chamber. Teasurement at any φ -plan can be performed as the QHA is a symmetrical antenna. For practical reasons, the yz-plane of QHA is aligned to the horizon and the input port of the QFN will point to the ground. The antenna is positioned onto the measurement turntable so that 360° radiation pattern measurement can be performed. At the transmitter, a co-polarised circular polarised conical antenna is used and position accordingly. Figure 4.9(b) shows the measured radiation of the antenna and also shown in the figure are simulation results for the radiation pattern of the QHA with QFN and QHA without QFN. It can be observed that measured radiation pattern has the same cardioid shape as the simulation results. The measured boresight gain is 4.2 dBic. The simulated boresight gain of the QHA, wihout QFN and QHA, with QFN are 5.1 dBic and 4.6 dBic, respectively. The 65 4.2 Parasitic Loop 0 −30o 30o 0 −5 −60o Magnitude, dB 0o10 dBic 60o −10 −10 −20 −90o −15 90o −20 −20 S11 S21 −25 S31 2.2 2.4 2.6 Frequency, GHz (a) S-parameter (Port 1) 2.8 120o 0 −150o S41 −30 2 Meas. QHA with QFN −10 Sim. QHA with QFN Sim. QHA(without QFN) −120o 10 dBic 150o 180o 3 (b) Elevation radiation pattern (φ = 90°, 2.45 GHz) Figure 4.9 Measured S-parameter and radiation pattern of the reference QHA. use of QFN has caused a change in the gain value. There are losses due to the microstrip circuit such as dielectric, conductor and radiation losses. The radiation from the microstrip could also distort the radiation pattern to a certain degree. The measured boresight gain is about 0.3 dB lower than that obtained through simulation. This loss could be attributed to variation in the properties in the substrate materials as well as imperfection caused by the fabrication and construction processes. Furthermore, the variation is within the typical tolerance of the measurement equipment. The boresight axial ratio of the QHA (with QFN) is measured using rotating reference antenna technique, as described in [51]. It involves illuminating the QHA with a reference linear polarised horn antenna. The reference antenna is rotated at 10° interval and the gain is recorded. Even though a smaller angle interval might yield a more accurate result, it is not practical with setup that was available. The measured axial ratio is found to be approximately 0.4 dB. The axial ratio from the simulation result for QHA with and without QFN is 0.0 dB. It has been mentioned in the previous paragraph that the variation in the properties of the substrate material as well as the fabrication and construction processes could caused this deviation. Nevertheless, the measured axial ratio is low and acceptable for the purpose of this analysis. 4.2 Parasitic Loop A parasitic loop has been shown to improve the boresight gain of the QHA [137]. The energy from the radiating QHA will couple into the parasitic loop and a significant current 66 4.2 Parasitic Loop Figure 4.10 One meandered section of the PML. will be induced in the parasitic loop when its resonant frequency is close to the resonant frequency of the QHA. However, since a typical loop element resonates when its circumference is about one wavelength, the radius of a circular loop can potentially be larger than the radius of a typical QHA. For example, the wavelength at 2.44 GHz is 123.0 mm where the loop radius needs to be about 19.5 mm. This is about double the radius of the reference QHA in Section 4.1 (Rqha = 10 mm). A loop with radius smaller or the same size as the radius of the QHA is more desirable as it will minimise the overall size increment and ease the fabrication process. In this section, two loop element designs will be investigated namely Parasitic Meandered Loop (PML) and Parasitic Quadrifilar Helical Loop (PQHL). The radius of these structures are restricted to be the same as the radius of the reference QHA. They will also be fabricated using the same materials and techniques as the reference QHA. 4.2.1 Parasitic Meandered Loop (PML) A meandered loop can be represented by a series of meandered sections. The general structure of one meandered section is shown in Figure 4.10. Lmh and Lmv represent the length of the horizontal and vertical subsection, respectively. Wml is the width of the loop element. The total length of the horizontal subsection, Lmhs is 8/3Lmh . Since Nl meandered sections need to be connected in series to form the circumference of the loop, each meandered sec2πR tion will have horizontal length of Nlqha where Rqha is radius of the reference QHA. From here, the length of the horizontal subsection, Lmh can be calculated using Eq. 4.4. The total length of one meandered section, Lms is the sum of the the length of the horizontal and vertical subsections, as indicated in Eq. 4.5. The total length of the meandered loop, LPML can also be calculated by multiplying the total length of each meandered section, Lms by the 67 4.2 Parasitic Loop (a) Nl = 4 (b) Nl = 8 (c) Nl = 16 Figure 4.11 PML with different number of subsection, Nl number of section, Nl as indicated in Eq. 4.6. 3πRqha 4Nl (4.4) 2πRqha + 2Lmv Nl (4.5) Lmh = Lms = LPML = 2πRqha + 2Nl Lmv (4.6) Variation of the number of meandered section, Nl and the meandered loop vertical subsection, Lmv From Eq. (4.6) and for a fixed radius of 10 mm, it is apparent that the deficiency in the loop length to achieve resonance at the desired frequency can be compensated by the appropriate combination of the vertical subsection, Lmv and the number of meandered section, Nl . Figure 4.11 shows the Parasitic Meandered Loops (PML) with Nl of 4, 8 and 16. The width of the element, Wml is fixed to 1 mm. Each Lmv is varied in order to obtain the desired resonant frequency. Figure 4.12 shows the required Lmv of the PML to achieve resonance of frequencies between 1.8 GHz and 3.2 GHz. Also shown in the figure is the required radius, Rcl of the parasitic circular loop (PCL) for the same frequency range. From the figure, it can be seen that higher Nl requires shorter Lmv at any particular resonant frequency. In order to work as a director, the parasitic loops need have resonant frequency higher than that of the radiating antenna. As an example, the physical characteristics of these loops with resonant frequency 68 4.2 Parasitic Loop 35 Meandered Loop (Nl = 16) Meandered Loop (Nl = 8) MPL Lmv length, mm 30 Meandered Loop (Nl = 4) Circular Loop(Radius) 25 20 15 10 5 0 1.8 2 2.2 2.4 2.6 2.8 Resonant frequency, GHz 3 3.2 Figure 4.12 PML resonant frequency. Table 4.2 Physical characteristics of the PML with different number of sections, Nl and reference circular loop (Resonant frequency of 2.80 GHz). Nl (mm) Lmv (mm) Lms (mm) 4 8 16 Circular Loop 9.4 7.00 5.75 18.70 (Radius) 138.03 174.83 246.83 117.50 (Circumference) 69 4.2 Parasitic Loop Figure 4.13 QHA with PML. of 2.80 GHz are summarized in Table 4.2. Even though higher Nl requires shorter Lmv , the total meandered loop length, Lms increases as Nl increases. The required Rcl for the PCL in order to resonate at 2.80 GHz is 18.7 mm which is still almost double the radius of the reference QHA. Figure 4.13 shows the placement of the PML on top of the reference QHA. The central core of the reference QHA is extended upwards in order to support the PML. The separation distance between the QHA and PML is Llg . The PCL is placed on top of the reference QHA. However, since the radius of the PCL differs from that of the QHA, the central core of the QHA cannot be used to support the PCL. If this antenna was to be fabricated, a supporting structure is needed to keep the PCL in place. However, in the simulation, the PCL is placed above the QHA without any support. In order to analyse the effect of PML resonant frequency on the gain of the QHA with different Nl values, a series of simulations are performed which involve varying Lmv while Llg is fixed to 12.5 mm. A similar set of simulations, using PCL with varying Rcl is also performed. Figure 4.14 shows the QHA gain over frequency with different parasitic loop configurations. In Figure 4.14(a), it can be seen that a PML (Nl = 4) with higher resonant frequency yields lower gain. The gain at 2.44 GHz increases from 5.9 dBic to 6.9 dBic as the PML resonant frequency decreases from 3.2 GHz to 2.6 GHz. This is about 0.8 dB to 1.8 dB improvement over the reference QHA. Using PML (Nl = 4) with resonant frequency of 3.0 GHz and 3.2 GHz, the gain improvement over the reference QHA is relatively consistent over the operating frequency range of 2.2 GHz to 2.7 GHz. However, at a resonant frequency of 2.8 GHz, there is almost no gain improvement compared wih operation at 2.7 GHz. This is despite it provides better gain im- 70 4.2 Parasitic Loop Lmv=10.8 mm (2.6 GHz) Lmv=8.3 mm (3.0 GHz) Ref. QHA Lmv=9.4 mm (2.8 GHz) Lmv=7.2 mm (3.2 GHz) 8 7 7 6 6 Gain, dBic Gain, dBic 8 5 4 3 3 2.3 2.4 2.5 Frequency, GHz 2.6 2 2.2 2.7 2.3 (a) PML Nl = 4 L =6.55 mm (2.6 GHz) mv Lmv=5.00 mm (3.0 GHz) Ref. QHA L =5.75 mm (2.8 GHz) mv Lmv=4.35 mm (3.2 GHz) 8 7 7 6 6 5 4 3 3 2.3 2.4 2.5 Frequency, GHz (c) PML Nl = 16 2.6 2.7 2.6 2.7 Rcl=20.00 mm (2.6 GHz) R =17.40 mm (3.0 GHz) cl Ref. QHA Rcl=18.70 mm (2.8 GHz) R =16.45 mm (3.2 GHz) cl 5 4 2 2.2 2.4 2.5 Frequency, GHz (b) PML Nl = 8 Gain, dBic Gain, dBic 8 Lmv=7.00 mm (2.8 GHz) Lmv=5.30 mm (3.2 GHz) 5 4 2 2.2 Lmv=7.95 mm (2.6 GHz) Lmv=6.00 mm (3.0 GHz) Ref. QHA 2 2.2 2.3 2.4 2.5 Frequency, GHz 2.6 2.7 (d) Parasitic Circular Loop Figure 4.14 Antenna gain over frequency with different parasitic loop (φ = 90°, θ = 0°). provement at 2.44 GHz operating frequency. When the PML resonant frequency is further reduced to 2.6 GHz, the QHA gain with PML is actually lower than the gain without PML above 2.5 GHz operating frequency. The resonant frequency of the PML is close or can be even lower than the operating frequency which causes the PML to act as a reflector. Similar characteristics, as described in the previous paragraph, can be observed in Figure 4.14(b) and Figure 4.14(c) for a PML with Nl of 8 and 16 sections, respectively. There seems to be only small variation in gain and with frequency between each PML. The QHA with PCL also exhibits the same general characteristics at operating frequency of 2.44 GHz, as can be seen in Figure 4.14(d). However, there is larger variation across the operating frequency range. The variation in gain with frequency between each PML and PCL can be attributed to their difference in impedance variation at each frequency point. The impedance 71 4.2 Parasitic Loop 0 7 −15 6 Gain, dBic Magnitude, dB −10 −20 Llg=2.5 mm −25 Llg=7.5 mm Llg=17.5 mm 5 4 Llg=7.5 mm −30 Llg=12.5 mm 3 Llg=17.5 mm −35 −40 2 Llg=2.5 mm Llg=12.5 mm 8 −5 Ref. QHA 2.2 2.4 2.6 Frequency, GHz 2.8 3 2 2.2 (a) Return Loss 2.3 2.4 2.5 Frequency, GHz 2.6 2.7 (b) Gain Figure 4.15 The effect of varying Llg for QHA with PML (Nl = 16, Lmv = 5.75 mm). variation causes differences in the relative phase between the radiation from each parasitic loop and radiation from the QHA. Variation of the distance between QHA and PML, Llg In the previous subsection, the separation distance between the QHA and PML, Llg was fixed at 12.5 mm. The relative phase variation between the radiation from the QHA and the coupled radiation from the parasitic loop is due to the change in the resonant frequency of the PML. The separation distance, Llg is another factor that could affect the relative phase between the two radiating signals, to a certain extent. For the purpose of this analysis, the PML with 16 subsections, Nl and vertical subsection length, Lmv of 5.75 mm is used. The resonant frequency of the PML is 2.80 GHz. Figure 4.15(a) shows the return loss when Llg is varied from 2.5 mm to 17.5 mm. The resonant frequency of the reference QHA without parasitic loop is 2.44 GHz. With a parasitic loop at 17.5 mm, 12.5 mm and 7.5 mm above the QHA, the resonant frequency drops to 2.435 GHz, 2.43 GHz and 2.41 GHz, respectively. When the separation distance is further reduced to 2.5 mm, the mutual coupling between the loop and the QHA starts to significantly affect resonant frequency causing it drops to 2.34 GHz. The effect of varying the separation distance, Llg , on the boresight gain can be seen in Figure 4.15(b). There is no significant effect on the peak gain with frequency when Llg is decreased from 17.5 mm to 7.5 mm where peak gain over frequency stays at about 6.9 dBic. However, due to the changes in resonant frequency when Llg is varied, the frequency at peak gain also vary slightly. However, when the separation distance is reduced to 2.5 mm, 72 4.2 Parasitic Loop 8 PML Nl=4 PML Nl=8 PCL Ref. QHA PML Nl=16 Gain, dBic 7 6 5 4 3 2 2.2 2.3 2.4 2.5 Frequency, GHz 2.6 2.7 Figure 4.16 Gain comparison between different parasitic loop configurations optimised for gain improvement for operating frequency between 2.35 GHz and 2.55 GHz. the peak gain drops to 6.6 dBic and occurs at 2.35 GHz. Optimised parameter for operating frequency between 2.35 GHz and 2.55 GHz. In the previous subsections, it has been demonstrated that there is a compromise between peak gain improvement and the range of operating frequency over which a reasonable gain improvement can be achieved by the PML. Although the PML could improve the gain by as much as 1.8 dB, the gain improvement is only occurs over a short frequency range. If the peak gain improvement is reduced by a small amount, a reasonable gain improvement could be achieved over a wider frequency range. The resonant frequency of the reference QHA is 2.44 GHz. The PMLs with different numbers of subsections, Nl are optimised to provide a reasonable levels of gain improvement at operating frequency between 2.35 GHz and 2.55 GHz. Figure 4.16 shows the gain comparison between different parasitic loop configurations optimised to provide gain improvement across the operating frequency range. The parasitic loops are configured such that they provide similar gain improvement over the operating frequency range. The physical characteristics for each parasitic loop are summarised in Table 4.3. The resonant frequency variation between the PML is small where the resonant frequency varies between 2.74 GHz and 2.80 GHz. The resonant frequency of the PCL is 2.9 GHz. All PMLs are capable of improving the gain by 1.6 dB, over the reference QHA at 2.44 GHz. The PCL has slightly better gain improvement where the improvement over the reference QHA is 1.8 dB. 73 4.2 Parasitic Loop Table 4.3 Physical characteristics of the parasitic loops optimised for gain improvement for operating frequency between 2.35 GHz and 2.55 GHz. Loop configuration Parameter (mm) Resonant Frequency (GHz) PML Nl = 4 PML Nl = 8 PML Nl = 16 PCL Lmv = 9.7 Lmv = 9.4 Lmv = 5.8 Rcl = 18 2.74 2.80 2.78 2.89 Radiation Pattern The radiation pattern of the QHA in the elevation plane (φ = 0°) with PML(Nl = 16, Lmv = 5.8) at 2.45 GHz is shown in Figure 4.17(a). Apart from improving the gain of the reference QHA by about 1.6 dB, the half power beamwidth (HPBW) is also decreased. Without the PML, the HPBW of the reference QHA is 125°. After incorporating the PML, the HPBW is reduced to 94°. The backside radiation for the QHA with and without the PML is −5.6 dB and −8.2 dB, respectively where it is dominated by radiation in the cross-polarisation state. The frontto-back gain ratio for the QHA with PML is 12.4 dB which is 1.1 dB lower that for the QHA without the PML. Figure 4.17(b) shows the elevation axial ratio (φ =0°) for the QHA with PML and the reference QHA. In general, both antennas perform similarly. The axial ratio below 3.0 dB spans between the elevation angle, θ of −77° and 77° as can be seen in Figure 4.17(b). This is beyond the HPBW of both QHA. The radiation pattern at 2.4 GHz, 2.45 GHz and 2.5 GHz is plotted in Figure 4.17(c). Apart from the change in boresight gain, no significant radiation pattern change can be observed. Prototype fabrication The parameters of the fabricated PML are summarised in Table 4.4. Figure 4.18 shows the PML together with the QHA and QFN. The PML is fabricated on the same substrate sheet as the QHA. This is one of the advantages of using PML with the same radius as the QHA. Both are then wrapped to form a cylindrical structure. Both ends of the PML are then soldered to form a complete loop. The radiation pattern and axial ratio were then measured using the same technique as discussed in Section 4.1. The elevation radiation pattern (φ =90°) is plotted in Figure 4.19. Also plotted are the radiation patterns from the antenna simulation with and without the QFN. It can be observed that the measured radiation pattern has the same cardioid shape 74 4.2 Parasitic Loop PML (Co−pol.) Ref. QHA (Co−pol.) −30o PML (Cross−pol.) Ref. QHA (Cross−pol.) 0o10 dBic Ref. QHA 30o 4 0 60o −10 Axial ratio, dB −60o PML 5 −20 −90o 90o −20 −10 −120o 3 2 1 120o 0 −150o 10 dBic 150o 0 −50 180o (a) Radiation pattern at 2.45 GHz 0 Elevation angle , (θo) 50 (b) Axial ratio at 2.45 GHz −30o 0o10 dBic 30o 0 −60o 60o −10 −20 −90o 90o −20 2.40 GHz −10 2.45 GHz 2.50 0 GHz −120o −150o 10 dBic 120o 150o 180o (c) Radiation pattern over frequency Figure 4.17 QHA with PML(Nl = 16, Lmv = 5.8) elevation radiation pattern and axial ratio at φ =0°. 75 4.2 Parasitic Loop Table 4.4 Physical parameters of the fabricated PML. Nl 16 Lmv (mm) Wml (mm) Llg (mm) 5.8 1 12.5 Figure 4.18 Fabricated PML with QHA and quadrature feed network. as the simulated results. However, the use of QFN deteriorated the boresight gain. With and without the QFN, the simulated boresight gains are 6.4 dBic and 6.8 dBic, respectively. The measured boresight gain is 6.0 dBic. The drops in the boresight gain as seen here are consistent with the results for the reference QHA discussed in Section 4.1. The drop in the simulated boresight gain of the antenna with and without QFN is due to the losses in the QFN network. The drop in the boresight gain for the measured antenna compared to its simulated counterpart could be attributed to the variation in the construction material and fabrication process. The axial ratio at the boresight is measured to be 0.3 dB. This is 0.3 dB higher than the axial ratio for the simulated antenna with and without the QFN. The difference is consistent with measured axial ratio for the reference as discussed in Section 4.1. 4.2.2 Parasitic Quadrifilar helix Loop (PQHL) The second parasitic loop that will be analysed is called parasitic quadrifilar helix loop (PQHL). The PQHL is constructed from quadrifilar helical elements. It can also be visualised as a combination of two bifilar helical elements. One bifilar loop of the PQHL is 76 4.2 Parasitic Loop −30o 0o10 dBic 30o 0 −60o 60o −10 −20 −90o 90o −20 Meas.−10 with QFN Sim. with QFN Sim. without QFN 0 −120o −150o 10 dBic 120o 150o 180o Figure 4.19 Measured elevation radiation pattern of the fabricated QHA with PML (φ =90°). shown in Figure 4.20(a). The bottom and top ends of the helical elements are shorted to their counterpart. The second bifilar helical elements is rotated by 90° with respect to the first bifilar elements. A complete PQHL with radius, R pqhl is shown in Figure 4.20(b). Even though one bifilar loop could have been used to improve the antenna gain, a quadrifilar is capable of producing a better gain based on initial analysis which is not covered in this thesis. Lq−element = Lq−middle + Sq−top + Sq−bottom + 2R pqhl (4.7) Lq−bi f ilar = 2(Lq−middle + Sq−top + Sq−bottom + 2R pqhl ) (4.8) Lq−ax = Lq−middle sin α pqhe (4.9) The PQHL in the unwrapped form is shown in Figure 4.20(c). Each helical element consists of a middle element, Lq−middle , Sq−top , Sq−bottom and two radial elements each having a length equals to the radius of the PQHL, R pqhl . The total length of one helical element is given by Eq. (4.7). One bifilar helical loop is constructed from two helical elements with the total length given by Eq. (4.8). The axial length or height of the PQHL depends on Lq−middle and its pitch angle, α pqhe as given by Eq. (4.9). In general, the PQHL can take any parameter dimensions as long as the resonant frequency of the bifilar helical loop is close to the centre operating frequency of the active 77 4.2 Parasitic Loop (a) Wrapped parasitic bifilar helical loop (b) Wrapped PQHL (c) Unwrapped PQHL Figure 4.20 Parasitic Quadrifilar Helix Loop (PQHL). 78 4.2 Parasitic Loop Table 4.5 Fixed parameters of the PQHL. R pqhl (Rqha ) 10 mm Sq−top (Stop ) Sq−bottom (Sbottom ) α pqhe (αqha ) Wpqhe (Welement ) 0.5 mm 1 mm 41.19° 2 mm 200 Lq−middle Lq−bifilar Lq−ax Length, mm 150 100 50 0 1.8 2 2.2 2.4 2.6 2.8 Resonant frequency, GHz 3 3.2 Figure 4.21 PQHL parameter length over frequency. antenna. However, for the purpose of this analysis, the PQHL follows the parameter dimension of the reference QHA from Section 4.1 except for the PQHL middle element length, Lq−middle . R pqhl , Sq−top , Sq−bottom , α pqhe and Wpqhe are fixed to Rqha , Stop , Sbottom , αqha and Welement , respective. For reference, these parameter values are summarised in Table 4.5. The axial height of the PQHL depends on Lq−middle and it will change as Lq−middle changes. Variation of the PQHL middle section length, Lq−middle Similar to the parasitic meandered loop (PML), as discussed in Section 4.2.1, the relative phase between radiation from the QHA and reradiation from the PQHL is largely determined by the resonant frequency of the PQHL. In this analysis, the length of the middle element of the PQHL, Lq−middle is varied to change the resonant frequency of the PQHL. Other parameters are fixed, as discussed previously. The resonant frequency of the PQHL is estimated from the simulation of the bifilar helical loop. A small slit is cut on the bifilar helical loop and a discrete port is attached. The required Lq−middle to achieve PQHL resonance from 1.8 GHz to 3.2 GHz is plotted in Figure 4.21. The resulting bifilar helical loop length, Lq−bi f ilar and axial ratio height, Lq−ax are shown as well. In order to evaluate the effectiveness of the PQHL with different Lq−middle , the PQHL is positioned on top of the reference QHA with a separation distance, Llq as shown in Figure 4.22. Llq is fixed to 12.5 mm. The central core of the QHA is extended to support a 79 4.2 Parasitic Loop Figure 4.22 PQHL placement on top of the QHA. 9 8 7 Gain, dBic 6 5 4 3 2 1 0 2.3 Lq−mid.=40.5 mm (2.6 GHz) Lq−mid.=36.5 mm (2.8 GHz) L =33.0 mm (3.0 GHz) L =30.0 mm (3.2 GHz) q−mid. q−mid. Ref. QHA 2.4 2.5 Frequency, GHz 2.6 2.7 Figure 4.23 Antenna gain with different PQHL middle element length, Lq−middle . PQHL which having the same radius. Figure 4.23 shows the resulting antenna gain with PQHL resonant frequency of 2.6 GHz, 2.8 GHz, 3.0 GHz and 3.2 GHz. A rather similar behaviour as the PML can be observed here where a higher resonant frequency will provide lower gain improvement over wider frequency range. On the other hand, at lower resonant frequencies, the antenna yields higher gain improvement but over narrower frequency range. At 2.45 GHz, the PQHL with resonant frequency of 2.6 GHz, 2.8 GHz, 3.0 GHz and 3.2 GHz provides gain improvement over the reference QHA of 2.6 dB, 1.7 dB, 1.2 dB and 1.0 dB, respectively. In general, these improvements are higher than the improvement provided by the PML. There is no significant change in the resonant frequency of the antenna due to varying the length of the middle section of the PQHLs, Lq−middle . 80 0 7.5 −10 7 6.5 −20 Gain, dBic Magnitude, dB 4.2 Parasitic Loop −30 Llg=0.0 mm −40 Llg=2.5 mm Llg=12.5 mm 2.2 2.4 2.6 Frequency, GHz (a) Return Loss 2.8 Llg=0.0 mm Llg=2.5 mm Llg=7.5 mm Llg=12.5 mm 4.5 Llg=17.5 mm −60 2 5.5 5 Llg=7.5 mm −50 6 L =17.5 mm lg 3 4 2.2 2.3 2.4 2.5 Frequency, GHz 2.6 2.7 (b) Gain Figure 4.24 The effect of varying Llg for QHA with PQHL (Lq−middle = 38 mm). Variation of the separation distance between QHA and PQHL, Llg The second parameter that can affect the relative phase between the radiation from the QHA and reradiation from the PQHL is separation distance, Llg . In this analysis, the length of the middle section of the PQHL, Lq−middle is fixed to 38 mm while the separation distance is varied from 0.0 mm to 17.5 mm. Unlike the separation distance analysis for the parasitic meandered loop (PML), where it needed to be greater than 0.0 mm, the separation distance for the PQHL can be made 0.0 mm without shorting the QHA elements. This done by allowing 45° angular offset between the bottom tips of the PQHL elements and the top tip of the QHA top stub. Figure 4.24(a) shows the return loss plots at one of the QHA ports as the separation distance is varied. The resonant frequencies of the QHA are 2.405 GHz, 2.420 GHz, 2.434 GHz, 2.437 GHz and 2.439 GHz when the separation distances are 0.0 mm, 2.5 mm, 7.5 mm, 12.5 mm and 17.5 mm, respectively. The resonant frequency of the reference QHA is 2.444 GHz. The resonant frequency shift at separation distance above 12.5 mm is minimal. At 0.0 mm, the resonant frequency shifts by approximately 39 MHz. It has been discussed in Section 4.2.1 that the closer the parasitic loop is to the QHA, the higher the mutual coupling between them will be. However, the level of resonant frequency shift is lower with the PQHL. For example, using a PML at separation distance of 2.5 mm, the resonant frequency shift is about 100 MHz whereas using the PQHL at the same separation distance, the shift is only 20 MHz. However, since the height of the PQHL height is greater than that of the PML, the effective relative phase between the radiation from QHA and reradiation from the parasitic loops can be different, even for the same separation dis- 81 4.2 Parasitic Loop Figure 4.25 Nested PQHL. tance. The change to antenna gain for different separation distances is shown in Figure 4.24(b). It can be observed that there is no significant changes to the gain level. The gain variations are less than 0.5 dB. The change in the relative phase between the radiation from the QHA and reradiation from the PQHL due to the changes in the separation distance is not significant enough to have large impact on the resulting antenna gain. Nested PQHL The PQHL has the same pitch angle and radius of the QHA and it is possible to place the PQHL in nested configuration, as shown in Figure 4.25. Simulation has shown that a QHA with nested PQHL (Lq−middle = 38 mm) has a boresight gain of 6.4 dBi, which is 1.2 dB higher than that of the reference QHA at 2.45 GHz. However, as it is placed is very close to the QHA, the mutual coupling is very high. The impedance reduces to 32 Ω and the resonant frequency shifts up by 20 MHz compared to the reference QHA. While the shift in frequency can be easily fixed, the nested configuration also causes a reduction in impedance bandwidth to 121 MHz. Radiation Pattern Here, the radiation patterns of the QHA with PQHL are compared to those of the reference QHA. The parameters of the PQHL are summarised in Table 4.5. The length of the middle section of the PQHL, Lq−middle and the separation distance, Llg between the QHA and PQHL are set to 38 mm and 12.5 mm, respectively. The radiation pattern at 2.45 GHz is shown in Figure 4.26(a). The boresight gain of the QHA with PQHL is 7.15 dBic which is 82 4.2 Parasitic Loop PQHL (Co−pol.) Ref. QHA (Co−pol.) o PQHL (Cross−pol.) Ref. QHA (Cross−pol.) 0o10 dBic −30 5 PQHL 30 4 Axial ratio, dB 0 −60o Ref. QHA o 60o −10 −20 −90o 90o −20 2 1 −10 −120o 3 120o 0 −150o 10 dBic 0 150o −50 180o (a) Radiation pattern at 2.45 GHz 0 Elevation angle , (θo) 50 (b) Axial ratio at 2.45 GHz −30o 0o10 dBic 30o 0 −60o 60o −10 −20 o 90o −90 −20 −10 GHz 2.40 2.45 GHz 0 2.50 GHz −120o −150o 10 dBic 120o 150o 180o (c) Radiation pattern over frequency Figure 4.26 QHA with PQHL(Lq−middle =38 mm, Llg =12.5 mm) elevation radiation pattern and axial ratio at φ =0°. 2 dB higher than the reference QHA. Apart from that, the half power beamwidth (HPBW) is also decreased. Without the PQHL, the HPBW of the reference QHA is 125°. After incorporating the PQHL, the HPBW is reduced to 92°. The backward antenna gain for the QHA with and without the PQHL are very similar at −8.4 dBic and −8.2 dBic, respectively. The backward radiation is predominatly in the cross-polarisation state (RHCP). The front-to-back gain ratio for the QHA with PQHL is 15.6 dB which is 2.1 dB higher than QHA without the PML. Figure 4.26(b) shows the elevation axial ratio (φ =0°) for the QHA with PQHL and the reference QHA. The QHA with PQHL has slightly wider angular coverage for axial ratio below 3.0 dB where it spans between the elevation angle, θ of −81° and 81° as indicated in Figure 4.26(b). This is 8° wider than that of the reference QHA. The radiation patterns at 2.40 GHz, 2.45 GHz and 4.2 Parasitic Loop 83 Figure 4.27 Fabricated PQHL(Lq−middle =38 mm, Llg =12.5 mm) with QHA and quadrature feed network. 2.50 GHz are plotted in Figure 4.26(c). Apart from the change in boresight gain, no significant radiation pattern change can be observed. Prototype Fabrication A PQHL is fabricated with the parameters as summarised in Table 4.5 in order to validate the design. The length of the middle section of the PQHL, Lq−middle and the separation distance, Llg between the QHA and PQHL are chosen to be 38 mm and 12.5 mm, respectively. The QHA is excited using the quadrature feed network (QFN), as shown Figure 4.27. The radiation pattern and axial ratio were measured using the same technique as discussed in Section 4.1. The elevation radiation pattern (φ =90°) at 2.45 GHz is plotted in Figure 4.28. Also plotted are the radiation patterns from the antenna simulation with and without the QFN. It can be observed that the measured radiation pattern has the same cardioid shape as the simulated results. However, the use of QFN has caused deterioration of the boresight gain. With and without the QFN, the simulated boresight gains are 6.64 dBic and 7.15 dBic, respectively. The measured boresight gain is 6.20 dBic. The drops in boresight gain observed here are consistent with the reference QHA results discussed in Section 4.1. The drop in the simulated boresight gain of the antenna with and without QFN is due to the losses in the QFN network. The drop in the boresight gain for the measured antenna compared to its simulated counterpart could be attributed to the variation in the 84 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) −30o 0o10 dBic 30o 0 −60o 60o −10 −20 −90o 90o −20 Meas.−10 with QFN Sim. with QFN Sim. without QFN 0 −120o −150o 10 dBic 120o 150o 180o Figure 4.28 Measured elevation radiation pattern of the fabricated QHA with PQHL (2.45 GHz, φ =90°, Lq−middle =38 mm). construction material and fabrication process. The axial ratio (2.45 GHz) at the boresight is measured to be 0.3 dB. This is similar to the level for QHA with PML which is 0.3 dB higher than the axial ratio for the simulated antenna with and without the QFN. The difference between the simulation and measurement result is consistent with the measured axial ratio of 0.4 dB for the reference QHA as discussed in Section 4.1. This indicates that degradation in the axial ratio due to the fabricated QFN rather than due to the PQHL. 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) In Section 4.2, it was demonstrated that the parasitic loop is capable of improving the antenna gain by configuring the parasitic loop as a director. Using similar concept to the parasitic loop, the parasitic element discussed in this section will be used to reflect and tilt the main beam to the side. The parasitic element is called a Side Parasitic Quadrifilar Helix Element (SPQHE). In reference to Figure 4.29, the radiating QHA is placed at the origin while the SPQHE is placed on the y-axis with separation distance of Ls−distance from origin. The SPQHE is rotated by 45° on its central axis as compared to the positing of the reference QHA on its central axis. As can be seen in Figure 4.29, the SPQHE is based on quadrifilar helix element. For the purpose of this analysis, the SPQHE is designed to have the same construction and parameters as the reference QHA in Figure 4.1. The exceptions to this requirement are the helical middle element length and helical number of turn. The length of the middle section 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) 85 Figure 4.29 SPQHE with QHA. of the SPQHE, Ls−middle will be varied in order to control its resonant frequency. The pitch angle of the Ls−middle is set equal to that of the reference QHA, αqha . The variation of the Ls−middle will change the helical number of turns and axial height of the SPQHE since the pitch angle is fixed. The length of middle element of the SPQHE, Ls−middle , axial height, Ls−ax and number of turns, Nspqhe is given by Eq. (4.10), Eq. (4.11) and Eq. (4.12), respectively where Stop , Sbottom , αqha and Rqha are the parameters of the reference QHA summarised in Table 4.1. Lq−element = Ls−middle + Stop + Sbottom (4.10) Ls−ax = Ls−middle sin αqha (4.11) Nspqhe = Ls−middle 2πRqha (4.12) Variation of the SPQHE middle element length, Ls−middle The SPQHE middle element length is allowed to be varied in order to change the resonant frequency of the helices. In this analysis, the separation distance, Ls−distance between 86 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) Table 4.6 Maximum antenna gain and tilt angle, θtilt with different SPQHE middle element length, Ls−middle (2.45 GHz). Ls−middle (mm) fspqhe (GHz) Max. gain (dBic) Tilt angle, θtilt 105.0 95.0 88.0 83.5 81 75 70 2.0 2.2 2.4 2.5 2.6 2.8 3.0 5.38 5.76 6.25 6.44 6.07 4.84 4.90 −10° −15° −15° −25° −30° −30° −25° the SPQHE and reference QHA is fixed to 61 mm which is about half the wavelength at 2.45 GHz. The only factor that will change the relative phase between the radiation from the QHA and reradiation from the SPQHE is the resonant frequency of the SPQHE, fspqhe since the separation distance is fixed. Three important characteristics that need to be observed as fspqhe varies are maximum antenna gain, the tilt angle, θtilt at the maximum gain, and the radiation pattern. Table 4.6 shows the effect of varying the length of the middle element of the SPQHE, Ls−middle on the maximum antenna gain and radiation tilt angle, θtilt at operating frequency of 2.45 GHz. The Ls−middle is varied from 105 mm to 70 mm and as a result, the SPQHE resonant frequency, fspqhe changes from 2.2 GHz to 3.0 GHz. At the SPQHE resonant frequencies of 2.2 GHz and 4.0 GHz the achieved tilt is only between −10° and −15°. Maximum tilt angle of −30° is achieved at 2.6 GHz and 2.8 GHz. However, the maximum gain at 2.8 GHz is only 4.97 dBic whereas it is 6.25 dBic at 2.6 GHz. The SPQHE with resonant frequency of 2.5 GHz has the same Ls−middle as that of the QHA, Lmiddle . Essentially, the SPQHE has the same dimension as the the reference QHA. With this PQHE, the maximum gain of 6.45 dBic is achieved with tilt angle of −2.5°. One of the objectives of using a SPQHE is to improve the antenna coverage by changing its radiation direction towards certain directions while at the same time reducing the coverage at the opposite direction. Figure 4.30 shows the radiation pattern of the antenna with varying fspqhe . It can be seen in the figure that, for a SPQHE with resonant frequency of 2.0 GHz, 2.8 GHz and 3.0 GHz, there is no significant change to the radiation pattern as compared to that of the reference QHA. A more significant change can be seen been for a SPQHE with resonant frequency of 2.4 GHz and 2.6 GHz where there is gain improvement at antenna elevation angles of −30° and gain reduction at an elevation angle of around 30°. 87 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) 0o10 dBic −30o 30o 0o10 dBic −30o 5 −60o −60o o 60 0 −5 −90o −90o o 90 o 90 fspqhe−5= 2.6 GHz −5 fspqhe = 2.0 GHz fspqhe0= 2.2 GHz fspqhe = 2.4 GHz fspqhe = 2.8 GHz 0 120o −120o fspqhe = 3.0 GHz 5 10 dBic 120o 5 Ref. QHA Ref. QHA −150o o 60 0 −5 −120o 30o 5 150o 10 dBic −150o 180o 150o 180o (a) 2.0 GHz, 2.2 GHz and 2.4 GHz. (b) 2.6 GHz, 2.8 GHz and 3.0 GHz. Figure 4.30 Antenna radiation pattern (φ = 90°) with varying SPQHE resonant frequency, fspqhe . Table 4.7 Antenna gain and tilt angle with varying SPQHE and QHA separation distance, Ls−distance as a fraction of wavelength at 2.45 GHz (Ls−middle = 83.5mm, 2.45 GHz). Ls−distance 1/3λ (40.8 mm) (49.0 mm) 1/2λ (61.2 mm) 3/4λ (30.4 mm) λ (122.4 mm) Ref. QHA 2/5λ Max. gain (dBic) Tilt angle, θtilt 6.19 6.50 6.45 6.30 5.75 5.15 −44° −35° −25° −1° 6° 0° Variation of the separation distance, Ls−distance between the SPQHE and QHA. The separation distance, Ls−distance also plays an important role in determining the behaviour of QHA incorporating SPQHE. For the purpose of this analysis, the length of the middle element of the SPQHE, Ls−middle is fixed to 83.5 mm. With this configuration, the SPQHE has the same dimensions as the reference QHA. Table 4.7 shows the effect of separation distance on the maximum antenna gain and its tilt angle, θtilt . At larger separation distance, less energy from the radiating QHA is able to couple into the SPQHE. The lowest gain of 5.75 dBic is achieved when the separation distance is one wavelength or 122.4 mm at 2.45 GHz. The gain gradually increases to 6.50 dBic as the separation distance is decreased to 2/5λ . With separation distance of 1/3λ , the gain is lower, which is due to mismatched loss. This will be explained later when discussing the impact on the return loss. It can also be observed in Table 4.7 that the tilt angle, θtilt increases with shorter separa- 88 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) −30o 0o10 dBic 30o −30o 5 −60o −60o o 60 −5 −90o o 90 −90o o 90 Ls−distance −5 = 40.8 mm −5 Ls−distance = 30.4 mm Ls−distance = 49.0 mm 0 Ls−distance = 61.2 mm 120o 0 Ls−distance = 122.4 mm −120o 120o Ref. QHA 5 Ref. QHA 5 10 dBic o 60 0 −5 −150o 30o 5 0 −120o 0o10 dBic 150o 180o (a) 40.8 mm, 49.0 mm and 61.2 mm. −150o 10 dBic 150o 180o (b) 30.4 mm and 122.4 mm Figure 4.31 Antenna radiation pattern (φ = 90°) with varying SPQHE and QHA separation distance, Ls−distance (Ls−middle = 83.5mm, 2.45 GHz). tion distance. The radiation from the radiating QHA needs to travel through the separation distance, before it reaches the SPQHE and the coupled energy at the SPQHE is lagging in time as compared to the radiation from the QHA. With the SPQHE shorted to ground plane, the interface between the SPQHE and ground plane has a reflection coefficient of 1 180◦ . The coupled energy will go through total reflection but the phase is reversed. The combination of the delay due to the separation distance and phase reversal causes the phase of the reradiated energy from the SPQHE to be advanced compared to the energy from the radiating QHA. The closer the separation distance, the higher phase advance. This is similar to a phased array antenna where higher phase advance will result in higher tilt angle. Figure 4.31 shows radiation pattern with varying separation distance. It can be seen that when the separation distance is half wavelength (61.2 mm) and below, it provides a significant coverage improvement over the reference QHA. The effect of separation distance between the radiating QHA and SPQHE on the return loss of the antenna is shown in Figure 4.32. It can be seen that the return loss increases with a decrease in the separation distance especially below 1/2λ (61.2 mm). The impedance bandwidth of the antenna is severely affected when the separation distance 1/2λ (40.8 mm). This is also the reason for the lower gain improvement, as discussed previously for this separation distance. The phase difference will approach 180° as the separation distance approaches zero (or as close as physically possible). This will cause the reradiating signal from the SPQHE to partly cancel the signal of the radiating QHA. A separation distance of 2/5λ and above would be a suitable compromise in term of performance and compactness. 89 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) 0 Magnitude, dB −10 −20 Ls−distance = 40.8 mm −30 Ls−distance = 49.0 mm Ls−distance = 61.2 mm −40 Ls−distance = 122.4 mm Ref. QHA −50 2 2.2 2.4 2.6 Frequency, GHz 2.8 3 Figure 4.32 Antenna return loss with varying SPQHE and QHA separation distance, Ls−distance (Ls−middle = 83.5mm, 2.45 GHz). 0o 10 dBic −30o 30o 5 −60o 60o 0 −5 −90o 90o −5 2.40 GHz 0 2.45 GHz 2.50 GHz −120o 120o 5 −150o 10 dBic 150o 180o Figure 4.33 Elevation radiation pattern (φ = 90°) of the SPQHE at different frequencies (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ )). Tilt angle, θtilt over operating frequency In the previous subsection, as Ls−distance is increased, θtilt decreases when compared at the same frequency for each configuration. However, when Ls−distance is fixed while the operating frequency is increased, the separation distance, as a fraction of the operating wavelength increases and θtilt will decrease. This effect can be seen in Table 4.8. θtilt varies from −40° to −30° as the frequency is varied from 4.0 GHz to 5.0 GHz. Figure 4.33 shows the radiation pattern of the SPQHA. It can be seen that the sidelobe level decreases as the frequency drops from 5.0 GHz to 4.0 GHz. 90 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) Table 4.8 The SQPHE tilt angle, θtilt and the respective gain at different frequencies (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ )). Frequency (GHz) Tilt angle, θtilt Gain (dBic) 3.75 GHz 4.00 GHz 4.25 GHz 4.50 GHz 4.75 GHz 5.00 GHz 5.25 GHz −42° −40° −38° −35° −33° −30° −28° 5.39 6.00 6.38 6.50 6.51 6.33 6.05 0o 10 dBic o 30o −30 5 −60o 60o 0 −5 −90o 90o −5 SPQHE open circuit SPQHE shorted to ground 0 Ref. QHA −120o 120o 5 −150o 10 dBic 150o 180o Figure 4.34 Elevation radiation pattern (φ = 90°) of the open circuit SPQHE (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ ), 2.45 GHz). Switched SPQHE From the previous analysis, there is a fixed connection from the SPQHE to ground plane. Instead of having a fixed connection, switches can be used to either make or break connection to the ground plane. Figure 4.34 shows the radiation pattern when the SPQHE (Ls−middle = 83.5mm, Ls−distance = 49.0mm, 2.45 GHz) is open-circuited from ground plane. It is apparent that there is negligible change in the radiation pattern shape as compared to the reference QHA. The boresight gain of the open-circuited SPQHE is only 0.2 dB lower than that of the reference QHA. When the SPQHE has the same configuration as the radiating QHA, the switches can also be used to swap the connection to the signal source from the existing radiating QHA. As the role of as a radiating antenna has been swapped, the tilt will be at the opposite quadrant. This configuration is known as Switched Active Switch Parasitic Antenna (SASPA). With 91 4.3 Side Parasitic Quadrifilar Helix Element (SPQHE) 3 SPQHE shorted to ground SPQHE open circuit Relative gain, dB 2 1 0 −1 −2 −3 −4 −5 −50 0 Theta, θo 50 Figure 4.35 Relative gain over middle radiation for the elevation radiation pattern of the open circuit SPQHE (Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ ), 2.45 GHz). SASPA configuration, the radiation pattern could be changed. One of the drawbacks of using SASPA configuration is that during switching, there will be a connection loss which is not desirable for continuous communication. In order to avoid signal discontinuity due to the switching of the radiating QHA, another SPQHE can be added to the opposite side of the radiating QHA at the same separation distance as the existing SPQHE. The signal source is always connected to the radiating QHA in the middle. The switches are only used to make or break connection between the SPQHEs and the ground plane. This configuration is known as Fixed Active Switched Parasitic Antenna (FASPA). Prototype Fabrication A SPQHE has been fabricated and its radiation pattern measurement was performed in order to validate its characteristics. The fabricated SPQHE is shown in Figure 4.36. The SPQHE has the same parameters as the reference QHA, as summarised in Table 4.1. The QHA is excited with the quadrature feed network (QFN) as described in Section 4.1. The separation distance, Ls−distance between the SPQHE and the QHA is fixed to 49.0 mm which is about 2/5-wavelength at operating frequency of 2.45 GHz. The measurement technique, as described in Section 4.1, was performed to measure the radiations pattern of the fabricated antenna. The elevation radiation pattern (φ = 90°) at 2.45 GHz is plotted in Figure 4.37. Also plotted is the simulated radiation pattern for the antenna with and without the QFN. It is important to compare with these result so that any imperfections caused by the QFN could be understood. 92 4.4 Independent Feed QHA with Optimal Combining Figure 4.36 Fabricated (Ls−distance =49.0 mm). SPQHE with QHA and quadrature feed network From the plots, in Figure 4.37, it can be observed that the general radiation pattern of the measured antenna agrees well with the simulated radiation pattern with and without the QFN. The simulated maximum gain with and without QFN is very close at 6.40 dBic and 6.50 dBic, respectively. The measured maximum gain of the fabricated antenna is 5.82 dB which has a loss of about 0.6 dB compared to the simulated result with QFN. The measured radiation pattern tilt angle, θtilt at the maximum gain is 36° which close to the simulated for the antenna with QFN which has tilt angle of 39°. The simulated tilt angle without QFN is 35°. The difference between the measured result and its simulated counterpart, with QFN, could be largely attributed to the construction of the QFN and SPQHE. This is consistent with variation in the measurement result for the QHA and parasitics loops, as seen from Section 4.1 and Section 4.2, respectively. 4.4 Independent Feed QHA with Optimal Combining Each received signal of the helical elements of the QHA can be represented by a 4x1 matrix, x as shown in Eq. (4.13) M x = sd ud + ∑ smum + n (4.13) m=1 where , sk , ud , um , n and M are the desired signal, interfering signal, 4x1 desired channel coefficient matrix, 4x1 interfering channel coefficient matrix, 4x1 noise signal matrix and the number of interfering signal, respectively. The received signal at each of the QHA 93 4.4 Independent Feed QHA with Optimal Combining 0o 10 dBic o 30o −30 5 −60o 60o 0 −5 −90o 90o −5 Meas. with QFN Sim. 0with QFN Sim. without QFN −120o 120o 5 −150o 10 dBic 150o 180o Figure 4.37 Measured elevation radiation pattern of the fabricated SPQHE with QHA (2.45 GHz, φ =90°, Ls−middle = 83.5mm, Ls−distance = 49.0mm (2/5λ )). helices can be combined with different complex weightings in order to maximise the gain towards a desired direction. The complex weights will amplify and introduce phase delay differently to each of the signals from the helical elements. The calculation of the weight becomes a minimum mean square error (MMSE) cost function problem. It can be shown that the optimum weight is provided by Weiner solution as represented by Eq. (4.14) where Rxx is the correlation matrix of the input signals, x. wopt = R−1 xx ud (4.14) According to the antenna reciprocity theorem, the optimum weight can also be applied to the input signals of each helical element during transmission to tilt the peak of beam towards a desired direction. The analysis in this section deals with tilting the peak of the beam without the presence of any interfering signal. The linear gain and phase information of each of the helical elements at the desired direction are used as the desired channel coefficient, ud in Eq. (4.13) and Eq. (4.14). The complex conjugate of the calculated optimum weights are then applied to the input signal of each of the helical elements and the resulting radiation patterns were analysed. This analysis is performed in MATLAB and the weight is then exported to computer simulation package to verify the radiation patterns. Based on the QHA, described in the previous section, the relationship between the achieved tilt angle and desired tilt angle, after the optimum the weights have been applied is shown Figure 4.38(a). It can be seen that there is a large tracking error between the desired and achieved tilt angle. The maximum achieved tilt angle is 20° which might not be 94 4.4 Independent Feed QHA with Optimal Combining 20 Achieved Tilt Angle (°) 5 4 3 2 1 0 0 Achieved Gain Axial Ratio 5 10 15 Achieved Tilt Angle (°) 15 10 5 0 0 20 (a) Achieved tilt angle vs. Desired tilt angle 20 40 60 Desired Tilt Angle (°) Achieved Tilt:0° Achieved Tilt:12° Achieved Tilt:18° 5 4 3 2 1 0 0 20 80 (b) Gain vs. Achieved tilt angle 6 Gain (dB) Achieved Gain and Axial Ratio (dB) 6 40 Theta (°) 60 80 (c) Radiation pattern at different achieved tilt angle Figure 4.38 Optimum combining result 100 4.5 Summary 95 sufficient for wide femtocell coverage. If the desired tilt angle is increased beyond 90°, the maximum gain occurs at the lower hemisphere. A plot of maximum gain against the achieved tilt angle is shown in Figure 4.38(b) as the tilt angle is increased from 0° to 16°. The maximum gain drops very gradually from 5.1 dB to 4.93 dB. However, as the achieved tilt angle progresses to 20°, the gain drop drastically to about 2.68 dB. Similar observation could be made for the axial ratio. The axial ratio remains below 1 dB up to tilt angle of 16° and increase dramatically to 4.6 dB as the tilt angle increases to 20°. There is a very good agreement between the result in MATLAB and computer simulation package. The radiation patterns on the φ = 0°-axis for the achieved tilt angle of 0°, 12° and 18° is shown in Figure 4.38(c). Titling the radiation pattern of the QHA will help in improving the average coverage of a small cell network. As an example, the curve of the 0° tilt intersects with the curve of 12° tilt at θ of 14°. Beyond this intersection point, tilting the radiation to 12° will have greater gain. However, as the gain decreases with increase in tilt angle, tilting the radiation to 18° will only benefit the coverage beyond θ of 56° as compared to tilting of 12°. 4.5 Summary In this chapter, the radiation mechanism and beamsteering capability of the QHA have been analysed. Due to the quadrature offset of each antenna element on the circumference of the QHA, the phase difference between elements is 90°. In order to combine the radiation of all element coherently, the QHA is fed in phase quadrature. It is capable of radiating a high quality circular polarised signal over wide beamwidth with excellent axial ratio. Two parasitic loops for QHA gain improvement has been successfully designed, fabricated and validated. The first design is based on parasitic meandered loop while the second design is based on parasitic quadrifilar helix loop. Both designs are capable of producing up to 1.8 dB gain improvement in the boresight direction. The advantage of this design is that it minimise the increase in volume by restricting the parasitic loop radius to the same size as the radiating QHA. Another parasitic element based QHA has been designed and evaluated. By using side parasitic quadrifilar element (SPQHE) at the side of the QHA, it gives the QHA the ability to tilt the beam to the side. The parasitic elements coupled the radiation from the radiating QHA and reradiated. The radiation from the SPQHE combined with radiation from the radiating QHA and as a result, the combined radiation is steered away from the PQHE. A 4.5 Summary 96 separation distance of about 0.4λ has been found to be a good comprise in steered angle and return loss level. Even though smaller closer separation distance can tilts the beam more, it is at the expense of lower gain due to the increase in the return loss. Finally, independent feed QHA is has been evaluated. Each of the antenna elements can be fed independently and optimal combining technique has been employed to steer the beam towards a certain direction. It has been shown that it is possible to steer the beam by as much as 16° without signification degradation on the maximum gain and axial ratio. Chapter 5 Side Parasitic and MIMO QHA Measurement Setup Public places such as shopping mall, airport and train station are suitable for small cell network deployment. Such places have high user density that demand for high data rate internet traffic. The demand can be met by deploying many short range base stations such as femtocells around the area. Each base station will serve a relatively small number of users. A photograph in Figure 2.1 shows the London Waterloo Station during a busy period. It can be seen that the users are distributed into clusters. The majority of the users are concentrated in front of the electronic schedule boards, inside the restaurants and at the benches. This kind of user distribution is typical of that found in other public places. If the base stations are equipped with antennas capable of beamsteering, they can work cooperatively to direct the beam toward each cluster. This will help to improve the average SNR between the BSs and MSs. Furthermore, co-channel interference can also be mitigated by avoiding two or more BSs covering the same cluster. In this chapter, a switched parasitic QHA (SPQHA) capable of beamsteering at the base station will be evaluated. The detail of these antennas will be discussed in Section 5.2. In this field measurement campaign, there are two base stations and each base station has a SPQHA with radiation pattern pointing to the middle and tilt to the left and right. 5.1 Measurement Site In the first measurement campaign (Section 3.4), the measurement was conducted at the concourse area within the School of Management building at University of Surrey, Guildford. However, the site is rather small for a two base stations measurement and due to this, 5.2 Antenna Configuration 98 the second measurement campaign were conducted at the plaza in front of the School of Management building, as shown in Figure 5.1(a). The area at the front and to the right of the plaza area were clear from any major obstruction for more than 70 m. To left of the plaza, there were a building with large glass windows about 30 m away. It was assumed that this building will not be a major signal reflector. It was expected that the reflection from the concrete floor to be more significant. The base stations are adjacent to the School of Management building wall and facing the plaza as can be seen in Figure 5.1(a). The first base station, BSA is situated on the left while the second base station, BSB is situated on the right when facing the two base stations. The channel sounder transmitter unit is placed close to BSB . The separation distance between the base stations is 6 m. It was powered by main supply from inside the building. The receiver unit is placed further away to the right of the plaza so that it will not cause obstruction during measurement. A long power cable was not convenient since it is further away from the building, and as such, it was powered by battery. The whole duration of the measurement campaign was video recorded. The view from the video camera can be seen in Figure 5.1(b). The video recording is used to determine the relative orientation of the users’ body and mobile stations with respect to the base stations. On the plaza floor, markers are placed at 1 m intervals. From these markers, a 15 m by 20 m virtual grid is superimposed onto the video recording as shown Figure 5.1(b). The grid helps to determine the relative position of the mobile stations during measurement. From the position information, the distance and angular position with respect to the base stations could also be determined. 5.2 5.2.1 Antenna Configuration Transmit Antennas Altogether, there are 18 transmit branches connected to the channel sounder transmitter unit. BSA and BSB have 7 and 11 antennas, respectively. The positioning of these antennas on the base stations’ masts are shown in Figure 5.2. In reference to Figure 5.2(a), BSA−le f t , BSA−mid and BSA−right form the three states of the parasitics antennas on BSA . Another four antennas elements is provided by MIMOA1 which has 4-independent inputs. In Figure 5.2(b), a similar configuration can be seen. BSB−le f t , BSB−mid and BSB−right form the three states of the parasitics antennas on BSB . In contrast to BSA , BSB has two 4-independent input MIMO antenna as represented by MIMOB1 and MIMOB1 . 99 5.2 Antenna Configuration (a) Without grid (b) With grid Figure 5.1 Measurement site. 100 5.2 Antenna Configuration (a) BSA (b) BSB Figure 5.2 Base stations’ antennas The height between the floor to the bottom antennas is approximately 2.4 m. The masts are down-tilted by about 25° so that the measurement range is within the half power beam width (HPBW) of the antennas. All transmit antennas are constructed based on the quadrifilar helix antenna (QHA) as discussed in the Section 4.1 with the parameter as summarised in Table 4.1. These antennas can be divided into two groups as will be discussed in the subsequent subsections. Side Parasitic Quadrifilar Helix Antenna (SPQHA) In Section 4.3, a Fixed Active Switched Parasitic Antenna (FASPA) based on the quadrifilar helix antenna (QHA) and side parasitic quadrifilar helix element (SPQHE) has been discussed. This antenna will be referred as Side Parasitic Quadrifilar Helix Antenna (SPQHA). The SPQHA can be an be constructed from one radiating QHA with one side parasitic quadrifilar helix element (SPQHE) on each side of the radiating QHA as discussed, in Section 4.3. The SPQHA has three different radiation patterns namely right, middle and left. The SPQHE connection to the its ground plane is controlled by RF switches. When the switches for the left and right SPQHE are opened, the SPQHA radiates to the middle. When the switch for the right (left) is closed, the SPQHA’s radiation tilted to the left (right). However, since the channel sounder is not capable of controlling the SPQHE switches, three separate antennas are needed to provide the three radiation patterns simultaneously, which can then be directly compared in post processing. In reference to Figure 5.2, the first group of the transmit antenna consists of the top 101 5.2 Antenna Configuration 0o 10 dBic −30o 30o 2 0 60o −10 −20 −90o 90o −20 Relative gain, dB −60o 0 −2 −4 −10 −120o −6 0BS B−left −150o 10 dBic 150o o 180 (a) Elevation radiation pattern φ = 90°) BSB−mid BSB−left 120o BSB−mid BSB−right (mirror) −50 0 Theta, θo 50 (b) Relative gain over BSB−mid Figure 5.3 SPQHA (BSB ) radiation pattern. three antennas at each base station. They are BSA−le f t , BSA−mid and BSA−right for BSA . On BSB , the antennas are BSB−le f t , BSB−mid and BSB−right . The subscript mid, le f t and right represent the direction or tilt of the antennas’ radiation. Each set of antennas represents one side parasitic QHA (SPQHA) in FASPA configuration. The SPQHE has the same physical construction as the radiating QHA and it acts as a reflector. Each antenna is fed using a quadrature feed network (QFN). The QFN used in the measurement campaign has a smaller footprint compared to the one designed in Section 4.1. This is achieved by using a substrate material with dielectric constant of 10. Ideally, a short separation distance between the these antennas on one base station is desirable in order to minimise the spatial diversity gain which would not be present when using a real switched parasitic antenna. However, when the antennas are placed closer to each other, their mutual coupling will rise. The resonant frequency and radiation pattern of the antennas might be disturbed and due this reason, the antennas are vertically separated by about one wavelength at 2.47 GHz. At this separation distance, the mutual coupling is less than −30 dB. Figure 5.3(a) shows radiation pattern of SPQHA on BSB . The BSB−mid have maximum gain of 4.7 dBic at the boresight direction. Figure 5.3(b) shows the measured relative gain of BSB−le f t over BSB − mid. The relative gain measurement is obtained from the elevation radiation pattern at φ = 90°and due to this, the radiation to the left and right are represented by −θ °-angle and +θ °-angle, respectively. In reference to the plot for BSB−le f t , it can be seen that below 16°, BSB−mid is stronger than BSB−le f t . However, above 16°, BSB−le f t is stronger than BSB−mid . Beyond 50°, BSB−le f t is about 2 dB stronger than BSB−mid . Similar 102 5.2 Antenna Configuration (a) MSA (b) MSB Figure 5.4 Mobile stations. behaviour can be observed for BSB−right where the radiation is tilted to the right. BSA−le f t , BSA−mid and BSA−right also have similar characteristic as previously discussed. QHA-based MIMO The second group of the transmit antennas consists of the bottom row antennas at each base station as can be seen in Figure 5.2. These antennas have been labelled as MIMOA1 , MIMOB1 and MIMOB2 . MIMOA1 is located at BSA while MIMOB1 and MIMOB2 are located at BSB . They are based on the independent input QHA where each set has four inputs. 5.2.2 Receive Antennas In this measurement campaign, four types of receivers are used. These receivers will be referred as Mobile Station A (MSA ), Mobile Station B (MSB ), Mobile Station B (MSA ) and omnidirectional antenna (Omnirx ). MSA and MSB are two-antenna receivers. MSC has four monopole antennas. However, the result for MSC will not be covered in this work. The channel measurements using MSA , MSB and MSC are performed simultaneously. Altogether, there are eight receive branches. In certain measurements, Omnirx which has a single antenna is used. When using this antenna, the connection to one of the antenna of MSA is disconnected and it is used for Omnirx . 103 5.2 Antenna Configuration 0 Magnitude, dB −5 −10 −15 −20 MSA1 MSA2 −25 2 2.2 2.4 2.6 Frequency, GHz 2.8 3 Figure 5.5 MSA return loss. 0o 0o 10 dB −30o 10 dB 30o −30o 0 −60o 60o −10 −60o −20 −90o 90o −90o 90o −20 −20 −10 −10 −120o 120o 10 dB −120o 120o Vertical pol. Horizontal pol. 0 −150 60o −10 −20 o 30o 0 o 150 Vertical pol. Horizontal pol. 0 o −150 o 10 dB 150o o 180 180 (a) Azimuth(θ = 90°) (b) Elevation(φ = 0°) Figure 5.6 Radiation pattern for MSA2 . Mobile Station A There are two antennas in the Mobile Stations A, MSA as indicated by MSA1 and MSA2 in Figure 5.4(a). Both antennas are designed based on meandered wired antenna. The measured return loss of these antennas are shown in Figure 5.5. The resonant frequency of MSA1 and MSA2 is 2.46 GHz and 2.47 GHz, respectively. Both antennas have approximately the same bandwidth 110 MHz which spans between 2.42 GHz and 2.53 GHz. Not shown in the figure is the top cover for the mobile station. Together with blue foam as the bottom cover as shown in the figure, they act as a protective casing for the fragile antennas. Figure 5.6 shows the radiation pattern for MSA2 . From the azimuth radiation pattern plot, the radiation to the right (φ = 90°, θ = 90°) is predominantly vertically polarised (VP) 5.2 Antenna Configuration 104 Figure 5.7 Omnidirectional antenna, Omnirx . with gain of approximately −0.1 dBi. The horizontal polarisation (HP) is approximately 10 dB lower. On the elevation plane (Figure 5.6(b), φ = 0°), the polarisation is largely horizontal with gain of approximately 0.2 dBi in most of direction. The radiation pattern for MSA1 is very similar to MSA2 with the exception that the radiation pattern is mirrored by the xz-plane. Mobile Station B Mobile Station B, MSB is constructed from the MS1 and MS2 as described in Section 3.1. MS1 and MS2 will be referred as MSB1 and MSB2 , respectively. These mobile stations are combined to form a two-antenna mobile station as shown in Figure 5.4(b). MSB1 has a meandered wire antenna and it is largely horizontally polarised in the upright orientation. As for MSB2 , it has a planar inverted-F antenna (PIFA). The dominant polarisation in the upright orientation is vertical polarisation. The radiation pattern for MSB1 (MS1 ) and MSB2 (MS2 ) can be found in Figure 3.3 and Figure 3.4, respectively. Omnidirectional Receive Antenna Shown in Figure 5.7 is the omnidirectional receive antenna, Omnirx . This antenna is manufactured by Electro-Metrics with model number EM-6865. It is vertically polarised in the azimuth plane (θ = 90°) with a typical gain of 0 dBi. 5.3 Measurement Equipment 5.2.3 105 Cable Configurations Ideally, the cable connecting the transmit and receive antennas to the channel sounder should have the same specifications such as length and loss. However, five different coaxial cables are used due to limited cable availability . The cable are arbitrarily named A, B, C, D and E to distinguish among them. The cable configuration is summarised in Table 5.1 together with the total loss and length. Most connections only need one cable. However, cables D and E are relatively heavy and bulky. They are not suitable to be connected directly to the antennas or channel sounder units. Cable A are used together with cables D and E as indicated in the summary table. The loss within each cable is compensated accordingly in post-processing. 5.3 5.3.1 Measurement Equipment Elektrobit Propsound Wideband MIMO Channel Sounder In this measurement campaign, the Elektrobit Propsound wideband MIMO channel sounder is used to probe the channel and record the measurement data. This is the same equipment as was briefly described in Section 3.4. Its operation and measurement setup will be discussed in more detail here. The channel sounder consists of separate transmitter and receiver units, as shown in Figure 5.8. They can be powered directly from the main supply or by batteries for mobile operation. The block diagram of the channel sounder transmitter and receiver units are shown in Figure 5.9(a) and Figure 5.9(b), respectively. The channel sounder is a direct-sequence spread spectrum system which uses BPSK modulated psuedo-noise (PN) code. Each PN code is chosen to have a good autocorrelation property. The modulated signal is then upconverted to the desired frequency before transmission. At the receiver, the received signal down-converted and demodulated. The signal is sampled and the raw I/Q-data is stored on the hard disk. In the post-processing stage, the impulse response of the channel is extracted from the raw I/Q-data using sliding window cross-correlator technique. The transmitter and receiver units of channel sounder only have one RF chain each and the connection to the array of antennas is controlled by the antenna switching unit (ASU). The channel sounder utilises time division multiplexing (TDM) to sequentially switched between the antennas. It is capable of capturing by up to 32x56 MIMO channels with null-to-null banwidth of 200 MHz. The channel sounder utilises a fast switching technique through Time Division Multiplexing (TDM). The acquisition period for each channel matrix 106 5.3 Measurement Equipment Table 5.1 Transmitters’ and receivers’ cables. Antenna Tx Rx BSA−le f t BSA−mid BSA−right MIMOA1−1 MIMOA1−2 MIMOA1−3 MIMOA1−4 BSB−le f t BSB−mid BSB−right MIMOB1−1 MIMOB1−2 MIMOB1−3 MIMOB1−4 MIMOB2−1 MIMOB2−2 MIMOB2−3 MIMOB2−4 MSA1 MSA2 MSB1 MSB2 MSC1 MSC2 MSC3 MSC4 Cable Type Total Loss(dB) Total Length(m) B 14.7 10 C 7.5 15 A,D,A 3.2 7 A 0.5 1 A,E,A 3.3 13 A,D,A 2.2 7 107 5.3 Measurement Equipment Figure 5.8 Elektrobit Propsound wideband MIMO channel sounder. needs to be less than the coherent time of the channel, Tc so that the channel could be considered constant [33]. The reference frequency in the transmitter and receiver units need to be synchronised in order to minimise phase rotation in the captured signal. There is a stable Rubidium based frequency clock for reference in the transmitter and receivers units. By linking these Rubidium clocks during initial setup, their frequency and phase tuning are adjusted to achieve synchronisation. Once they are synchronised, they can be unlinked and the transmitter and receiver units can be separated. The Rubidium clocks have backup power and as such the transmitter and receiver units can be switched off during transport. 5.3.2 Equipment Parameter In this measurement campaign, the measurement is performed at centre frequency, fc of 2.47 GHz with bandwidth of 50 MHz. The transmit power is set to 10 dBm. The maximum walking speed, vwalk−max for the user holding the mobiles stations is limited to about 1 m s−1 . The maximum Doppler shift, fd−max can be calculated from Eq. ( 5.1) fd−max = vwalk−max fc c (5.1) where c is the speed of light. In this measurement campaign, the maximum Doppler is shift is 6.59 Hz. According to the Nyquist sampling criteria, the sampling frequency need to be 108 5.3 Measurement Equipment (a) Transmitter unit (b) Receiver Unit Figure 5.9 Block diagram of the Elektrobit Propsound wideband MIMO channel sounder. 109 5.4 WLAN Interference at least twice the maximum Doppler shift so that the signal will be accurately sampled. In order to meet this criteria, the sampling frequency, fs for this measurement campaign is set to 21.57 Hz. There are 18 transmitters and 8 receivers combination as discussed in Section 5.2. The channel measurement is started by transmitting from the first antenna, T x1 . At the receiver, the data is sequentially recorded for the Rx1 to Rx8 . Before switching the transmission to the second transmitter, T x2 , one guard transmission using T x1 is performed. This measurement cycle is repeated until all transmitters and receivers combination has been covered. In total, one acquisition cycle contains 162 coded transmissions consisting of 18x8 data codes and 18 guard codes. In order to ensure that the 162 coded transmissions experience constant or the same propagation channel condition within one acquisition cycle, the acquisition cycle need to be completed within the coherence time of the channel, TC . The coherence time of the channel is 31.5 ms as estimated by Eq. (5.2). The time duration of the tranmission code depends on the chip duration and number of chips. The chip rate is half the bandwidth of the channel measurement. The chip rate is 25 MHz with two samples per chip since the bandwidth is set to 50 MHz. The chip duration is 20 ns. The transmission code length is set to 1022 chips and each code duration is 20.4 µs. With 162 codes, the duration of each acquisition cycle is 3.3 ms. The channel can be considered constant as the acquisition cycle time is less than the coherence time of the channel. TC = 9 16π fd−max (5.2) The measurement and equipment parameters for the measurement campaign is summarised in Table 5.2. 5.4 WLAN Interference Part of the measurement frequency range coincides with the frequency of the WLAN, and due to this, there were instances where the captured signal was interfered by the WLAN signal. In Section 3.4, the interference was removed from the narrowband channel data. It is only applicable for that particular sample frequency. In this section, the interpolation was performed to the power delay profile (PDP) of the channel. Initially, the noise floor of the data without interference was established. The PDP instance with interference could 110 5.5 Measurement Scenarios Table 5.2 Measurement and equipment parameters. Parameter Carrier frequency, fc Measurement bandwidth Transmit power Number of transmit branch Number of receive branch BS height BS separation distance Maximum MS speed, vwalk−max Sampling frequency, fs Code length Chip duration Value 2.47 GHz 50 MHz 10 dBm 18 8 2.4 m 6m 0.8 m s−1 21.57 Hz 1022 20 ns be detected where its noise floor will rise relative to the noise floor without interference. If the noise floor with interference rose above a certain threshold, that PDP instance was discarded and replaced with interpolated data using clean samples before and after that particular sample. The longest sample sequence with interference is less than seven and the total interpolated sample in each measurement is less than 6 %. It is expected that interpolated samples will not cause significant distribution deviation from the true distribution of the samples. 5.5 Measurement Scenarios Figure 5.10 shows the zones and paths for the movement of the mobile stations during different measurement scenarios. Path A represents a straight path in front of BSB while Path B represents the straight equidistant path between BSA and BSB . Zone A and Zone B cover the area in front of BSA and BSB , respectively. The mobile stations in these zones do not cross the middle path between BSA and BSB as indicated by Path B. On the other hand, in Zone C and Zone D, the mobile stations cross the middle path. There are nine measurement scenarios. The detail of each measurement scenario is described below and summarised in Table 5.3. MeasPL− f ront : Path loss measurement in front of BSB along Path A using Omnirx . Measurement is performed between 3 m to 18 m from BSB . Total duration is approximately 30 s with 600 data points. 5.5 Measurement Scenarios 111 Figure 5.10 Measurement paths and zones. MeasPL−mid : Path loss measurement in between of BSA and BSB along Path B using Omnirx . Measurement is performed between 3 m to 18 m from BSB . Total duration is approximately 30 s with 600 data points. MeasFar−data : MSA and MSB are randomly moving in Zone A and Zone B, respectively. For majority of time, the mobile stations are relatively far from each other. Both mobile stations are in Video Mode. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data points. MeasFar−talk : MSA and MSB are randomly moving in Zone A and Zone B, respectively. For majority of time, the mobile stations are relatively far from each other. Both mobile stations 5.5 Measurement Scenarios 112 are in Talk Mode. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data points. MeasNear−data : MSA and MSB are randomly moving in Zone C and Zone D, respectively. For majority of time, the mobile stations are relatively close to each other as they cross the middle path as indicated by Path B. Both mobile stations are in Video Mode. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data points. MeasNear−talk : MSA and MSB are randomly moving in Zone C and Zone D, respectively. For the majority of time, the mobile stations are relatively close to each other as they cross the middle path as indicated by Path B. Both mobile stations are in Talk Mode. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data points. MeasFar−talk/data : MSA and MSB are randomly moving in Zone A and Zone B, respectively. For the majority of time, the mobile stations are relatively far from each other. MSA and MSB are in Talk Mode and Video Mode, respectively. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data points. MeasNear−talk/data : MSA and MSB are randomly moving in Zone C and Zone D, respectively. For the majority of time, the mobile stations are relatively close to each other as they cross the middle path as indicated by Path B. MSA and MSB are in Talk Mode and Video Mode, respectively. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data points. MeasWalkaround : MSA and MSB are placed on a bench at about 1 m above ground. The benched is located at 5 m away and 15° to the right of BSB . Two users are randomly moving around the bench which will intermittently block the link between the MSs to BSs. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data 113 5.6 Post Processing Table 5.3 Measurement scenarios. Name MS: Zone (Mode) MeasPL− f ront MeasPL−mid Omnirx : Path A. Omnirx : Path B. MSA : Zone A (Video Mode). MSB : Zone B (Video Mode). MSA Zone A: (Talk Mode). MSB Zone B: (Talk Mode). MSA : Zone C (Video Mode). MSB : Zone D (Video Mode). MSA : Zone C (Talk Mode). MSB : Zone D (Talk Mode). MSA Zone A: (Talk Mode). MSB Zone B: (Video Mode). MSA : Zone C (Talk Mode). MSB : Zone D (Video Mode). MSA : All zones (Video Mode). MSB : All zones (Video Mode). MSC : Stationary MSA : Around MSC (Video Mode). MSB : Around MSC (Video Mode). MeasFar−data MeasFar−talk MeasNear−data MeasNear−talk MeasFar−talk/data MeasNear−talk/data MeasWalkaround MeasMSC Meas. Total count sample Total duration (s) 1 1 ~600 ~600 ~30 ~30 2 ~1200 ~60 2 ~1200 ~60 2 ~1200 ~60 2 ~1200 ~60 2 ~1200 ~60 2 ~1200 ~60 2 ~1200 ~60 2 ~1200 ~60 points. MeasMSC : MSC together with MSA and MSB are used in this measurement. MSC is stationary at approximately 8 m in front of BSB and 1 m above the ground. MSA and MSB are randomly moving in around MSC . MSA and MSB are in Video Mode. Measurements are performed twice with approximately 30 s in duration and 600 data points for each measurement. In total, the measurement duration is approximately 60 s with 1200 data points. The analysis of this measurement scenario is not covered in thesis. 5.6 Post Processing A back-to-back measurement from the transmitter to the receiver units is performed in the initial setup. This will compensate any loss or phase delay due to the equipment. In post pro- 5.7 Path Loss Compensation From The Measured Data 114 cessing, further compensation is need to take into account the transmit and receive antenna gain and cable loss. The power delay profile provide a wideband channel data. A narrowband data can be extracted from the wideband data by applying FFT operation. From the FFT result, narrowband data is usually extracted at the centre frequency. 5.7 Path Loss Compensation From The Measured Data In each measurement scenario, the MSs are randomly moving in the measurement zone. Depending on the location of MS, differences in the path loss between the MS to BSA and BSB can be significant. In order the evaluate the diversity gain, the the path loss needs to be compensated by post processing the data. It was discussed in Section 5.1 that, the location of the MSs with respect to the BSs can be estimated from the virtual grid in the video recording. The location of the MSs are determined at a certain interval and a special attention is given when the MSs changes direction. The path loss between two intervals is then interpolated. With this information, the path loss is compensated from the narrowband data. 5.8 Antenna Selection Combining In order to evaluate the diversity gain of using multiple antenna, antenna selection combining techniques are used. Using this technique, the system monitors the signal from all available branches simultaneously and selects the branches with the strongest signal. The diversity gain can be estimated from complementary cumulative density function (CCDF) plot at 99% probability level. It must be noted that a real switched parasitic antenna in FASPA configuration only has one RF chain and the signal from each parasitic state will be sequentially scanned at a certain interval. In this work, there are three antenna selection schemes that are evaluated as follows: Local Antenna Selection (LAS): Antenna selection is performed from branches available locally at each particular BS only. For example, BSA will perform antenna selection from BSA−le f t , BSA−mid and BSA−right while for BSB , it will perform antenna selection from BSB−le f t , BSB−mid and BSB−right . Middle Antenna Selection(MAS): The base stations are in cooperative mode and the an- 115 5.9 Result I: Diversity Gain 10 10 BSB−right 5 BSA−mid 5 BSB−mid 0 BSA−left 0 BSB−left Magnitude, dB Magnitude, dB BSA−right −5 −10 −15 −5 −10 −15 −20 −20 −25 −25 −30 10 15 20 25 Time, s 30 35 40 (a) BSA −30 10 15 20 25 Time, s 30 35 40 (b) BSB Figure 5.11 Local mean for MSA1 (MeasNear−data ). tenna selection is performed between middle radiating antennas which is BSA−mid and BSB−mid . Global Antenna Selection(GAS): The base stations are in cooperative mode. Six parasitic antennas branches are available for antenna selection operation. These antennas are BSA−le f t , BSA−mid and BSA−right , BSB−le f t , BSB−mid and BSB−right . 5.9 5.9.1 Result I: Diversity Gain Local mean diversity The local mean of the received signal can be obtained by averaging the signal over a short distance as provided by Eq. (3.3). This will remove the fast fading component from the received signal. From here, the antenna selection combining can be performed using the local mean signal. Using this technique, the gain due to antenna gain difference can be evaluated. Figure 5.11 shows the local mean signal for MSA1 from the MeasNear−data measurement. In this measurement, MSA is randomly moving in Zone C as indicated in Figure 5.10. MSA is located between BSA and BSB for majority of the time and it can be seen that BSA−right and BSB−le f t is the dominant signal. Even though there is high correlation between the signal on one base station, the signal means are different which is due to parasitic antenna gain. By performing antenna selection combining, the difference in then antenna gain can 116 0 0 50 50 80 80 Cumulative Probability Cumulative Probability 5.9 Result I: Diversity Gain 90 99 BSA−right BSA−mid 90 99 BSB−right BSB−mid BSA−left BSB−left LAS(BSA) 99.9 −35 −30 −25 −20 −15 −10 Magnitude, dB −5 0 LAS(BSB) 5 (a) BSA 99.9 −35 −30 −25 −20 −15 −10 Magnitude, dB −5 0 5 (b) BSB Figure 5.12 CCDF plot of the local mean for MSA1 at each base station (MeasNear−data ). be harvested. The CCDF plot shown in Figure 5.12 confirms this phenomenon. It can be seen that LAS improves the signal by approximately 1 to 2 dB throughout the signal range at both BSs. The CCDF plot in Figure 5.13 compares the diversity gain with different antenna selection groups. It has been mentioned in the previous discussion that, LAS provide diversity gain of approximately 2 dB relative to the middle parasitic antenna. By switching between the middle parasitic antenna (MAS), 3 dB diversity gain can be obtained. This is to be expected as when the MS changes direction, the body can be blocking one BS and have a clear link with the other base station or vice versa. A significant diversity gain by using GAS can be seen in the plot. Compared to the middle parasitic antenna, GAS provides approximately 5 dB diversity gain. 5.9.2 Instantaneous power diversity In this subsection, the diversity gain will be calculated from instantaneous received power which is consisting of the local mean as well as fast fading component. By using the instantaneous received power, the diversity gain due to antenna gain together with fast fading can be quantified. This analysis is peformed using the same data set as employed in the previous subsection (MeasNear−data ). The diversity gain with different antenna selection schemes can be compared from the the CCDF plot in Figure 5.14. LAS at either BSs yields similar diversity gain of approximately 8 dB relative to the middle parasitic antenna. In contrast to LAS on the local mean, 117 5.9 Result I: Diversity Gain 0 Cumulative Probability 50 80 90 BSA−mid 99 BSB−mid LAS(BSA) LAS(BSB) MAS GAS 99.9 −35 −30 −25 −20 −15 −10 Magnitude, dB −5 0 5 Figure 5.13 CCDF plot of the local mean for MSA1 at each base station in measurement MeasNear−data with GAS. Cumulative Probability 0 50 80 90 BSA−right BSA−mid BSA−left 99 BSB−right BSB−mid BSB−left LAS(BSA) 99.9 LAS(BSB) MAS GAS 99.99 −50 −40 −30 −20 −10 Magnitude, dB 0 10 Figure 5.14 CCDF plot of the received signal for MSA1 at each base station in measurement MeasNear−data with GAS. 5.9 Result I: Diversity Gain 118 this diversity gain is approximately 6 dB higher. This indicates that the additional diversity gain is due to fast fading signals. The effect of vertically separated antenna has been analysed in [172]. It has been shown that the correlation coefficient of signals from two antennas with vertical separation of one wavelength in open space environment could be higher than 0.8 High correlation coefficient leads to low diversity gain. In residential area, the correlation coefficient is less than 0.6 The difference between these environments could be attributed to the angle spread of the incoming signals. The angle spread in the open space environment and residential area are 8° and 20°, respectively. In this measurement campaign, the measurement was performed in open space environment and the separation distance between the parasitic antennas on one BS is approximately one wavelength at 2.47 GHz. In relation to the discussion in the previous paragraph, it could be expected that correlation coefficient between the parasitic antenna branches to be high which will lead to low spatial diversity gain. The diversity gain achieved using LAS indicates that, despite the close spacing between the parasitic antennas, the change in radiation pattern is capable of decorrelating between the received signals. The diversity gain due to the change in radiation pattern is known as pattern diversity. It could be concluded that the LAS diversity gain could be largely attributed to the pattern diversity as provided by the different states of the parasitic antennas. Coincidently, MAS also yield approximately 8 dB diversity gain. The decorrelation of the signals from both middle parasitic antennas is mainly due to the separation distance between the BSs. This diversity due to spatial separation is known as spatial diversity. Finally, GAS provides diversity gain of approximately 13 dB relative to the middle parasitic antenna. This is approximately 5 dB higher than MAS and LAS. The diversity gain could be attributed to the combination of spatial diversity and pattern diversity. Firstly, the spatial diversity is provided by switching between BSA and BSA which has large separation distance. Secondly, the pattern diversity comes from the different states of the parasitic antennas available at both BSs. In the previous analysis, the data set comes from MeasNear−data . Figure 5.15 shows the CCDF plot of the received signal for MeasNear−talk , MeasFar−data and MeasFar−talk . The subscript talk and data indicates the MS orientation as in Talk Mode and Data Mode, respectively. The diversity gain is summarised in Table 5.4. It can be seen that, the result agree well with each other. This suggest that the MS orientation and the location of the user in the cell do not play a significant role in term of diversity gain. So far, the diversity gain is analysed based on both BSs serving a single user. The 119 5.9 Result I: Diversity Gain Cumulative Probability 0 50 80 90 BSA−right BSA−mid BSA−left 99 BSB−right BSB−mid BSB−left LAS(BSA) 99.9 LAS(BSB) MAS GAS 99.99 −50 −40 −30 −20 −10 Magnitude, dB 0 10 0 50 0 50 80 90 80 90 Cumulative Probability Cumulative Probability (a) MeasNear−talk BSA−right BSA−mid BSA−left 99 BSB−right BSB−mid BSB−left LAS(BSA) 99.9 99.99 −50 −40 −30 −20 −10 Magnitude, dB BSA−right BSA−mid BSA−left 99 BSB−right BSB−mid BSB−left LAS(BSA) 99.9 LAS(BSB) LAS(BSB) MAS GAS MAS GAS 0 10 99.99 −50 −40 (b) MeasFar−data −30 −20 −10 Magnitude, dB 0 10 (c) MeasFar−talk Figure 5.15 CCDF plot of the received signal for MSA1 at each base station in measurement MeasNear−data with GAS. Table 5.4 Summary of instantaneous power diversity gain in different measurements relative to the middle parasitic antenna. Measurement LAS(BSA ) LAS(BSB ) MAS GAS MeasNear−data MeasNear−talk MeasFar−data MeasFar−talk 8 dB 9 dB 9 dB 9 dB 8 dB 9 dB 9 dB 9 dB 8 dB 8 dB 7 dB 9 dB 13 dB 13 dB 14 dB 14 dB 120 5.10 Result II: Evaluation of the Downlink Interference Cumulative Probability 0 50 80 90 BSA−mid 99 BSB−mid Primary user: LAS(BSA) Primary user: LAS(BSB) 99.9 Primary user: MAS Primary user: GAS Secondary user:GAS 99.99 −50 −40 −30 −20 −10 Magnitude, dB 0 10 Figure 5.16 CCDF plot of the received signal for MSA1 as secondary user (MeasNear−data ). antenna selection is performed based by providing priority for the primary user to select the BS. In MeasNear−data , apart from MSA , there is another MS which is MSB . If both BSs are serving MSB , MSA becomes the secondary user. It is assumed that MSA as the secondary user will be served by the base station that is not being used by the primary user, MSB . As for example, if MSB is the primary user, it has the priority to be served either by BSA or BSB . Once it has been determined which BS will be serving BSB , the unused BS will used to serve MSA . As the secondary user, MSA could not choose the BS and will always be served by the unused BS. Figure 5.16 shows the CCDF plot of the received signal for MSA as secondary user (black line) with GAS. It can be seen that there is approximately 3 dB diversity gain reduction compared to when MSA is the primary user with GAS. Even though there is deterioration of the diversity gain, it is not worse than MAS and LAS as a primary user. Even when both BSs are serving another user, the secondary user can still benefit from LAS on the BS that is not being used at that particular instance. 5.10 Result II: Evaluation of the Downlink Interference Due to scarcity of spectrum, base stations in a small cell network will typically utilise the same carrier frequency. This will give rise to co-channel interference (CCI). This situation can be severe especially close to the cell edge where the signal strength from the adjacent base stations can be at similar level. The signal-to-interference ratio (SIR) is defined as the 121 0 0 50 50 80 80 Cumulative Probability Cumulative Probability 5.10 Result II: Evaluation of the Downlink Interference 90 99 90 99 CASE 1 CASE 3 CASE 4 CASE 1 CASE 2 99.9 −40 −30 −20 −10 SIR, dB 0 10 20 99.9 −40 −30 (a) LAS −20 −10 SIR, dB 0 10 20 (b) GAS Figure 5.17 CCDF plot of the SIR for MSA1 (MeasNear−data ). Table 5.5 Summary of interference scenarios at MSA 1. Description CASE 1 The intended signal is from BSA−mid and interference is from BSB−mid . CASE 2 LAS is applied at BSA and BSB independently. The intended and interference signals are assumed to be from BSA and BSB , respectively. CASE 3 MAS is applied with priority given to MSA . CASE 4 GAS is applied with priority given to MSA . ratio between the power of intended signal, Pd to the power of the interference signal Pi as follows: SIR = Pd Pi (5.3) The interference and antenna selection scheme employed is summarised in Table 5.5. Local Antenna Selection In reference to Figure 5.17(a), CASE 1 plot refers to the SIR MSA1 with intended signal from BSA−mid and the interference from BSB−mid . When BSA and BSB are serving MSA and MSB with LAS, respectively, the SIR at MSA due to interference from BSB is represented by the CASE 2 plot. Despite the fact that no SIR improvement scheme has been deployed, it can be seen that, at 99% probability level, the SIR improved by approximately 7 dB. The 5.11 Result III: Comparison with QHA-based MIMO 122 SIR improvement can be partly attributed to diversity due to LAS at BSA . It is also possible that while BSB is serving MSB with LAS, the interference to MSA is reduced. Middle and Global Antenna Selection With MAS, BSA−mid and BSB−mid give priority to MSA . Its SIR is represented by CASE 3 plot in Figure 5.17(b). CASE 1 plot is the same CASE 1 scenario as in the previous subsection. It can be seen that there is a 20 dB SIR improvement compared to CASE 1. With GAS, both BSs give priority to MSA . The objective is to maximise the intended signal power and minimise the interference signal power. The strongest signal will be selected first, then the minimum signal from the other base station is selected. The SIR for this scenario is represented by CASE 4 plot in Figure 5.17(b). There is approximately 22 dB SIR improvement compared to CASE 1. Even though the SIR improvement is similar to CASE 3 at low signal level, there is generally about 3-5 dB improvement at high signal level. 5.11 Result III: Comparison with QHA-based MIMO In Section 5.2, it has been discussed that BSA and BSB have one and two 4-element QHAbased MIMO antennas, respectively. The antenna on BSA , it is referred to as MIMOA1 while on BSB as MIMOB1 and MIMOB2 . These antennas essentially have the same physical dimension as BSA−mid and BSB−mid . The only different is the feeding network. Instead of using a fixed quadrature feed network as in BSA−mid and BSB−mid , the QHA-based MIMO antennas use independent microstrip feed networks. The performance of this setup will be compared to that of switched parasitic QHA (SPQHA). In order to form the MIMO link between the BSs and MS, both receive antennas on the MSA will be used in the analysis with the data set obtained from MeasNear−data . When using a single QHA-based MIMO antenna on either BS, a 4x2 channel matrix is formed. If two QHA-based MIMO antennas are used either on BSB or cooperatively by BSA and BSB , a 8x2 channel matrix is formed. When using the SPQHA, one of the parasitic antennas on each BS is selected to form a enhanced 2x2 channel matrix at one time instance. This setup has more antennas than it is actually used and it is known as MIMO antenna selection [173]. The performance comparison will be based on eigenvalue analysis. A maximum of two eigenvalues can be obtained at one particular instance as the minimum number of transmit or receive antenna is two. The parasitic antenna selection criteria for the MIMO SPQHA will be based on two schemes. The first scheme is to maximise the second eigenvalue, λ2 123 5.11 Result III: Comparison with QHA-based MIMO and the second scheme is to maximise the first eigenvalue, λ1 [174]. In each measurement data set, the channel matrix is normalised according to NRx NT x 1 Nnorm = ∑ ∑ E{|hi j |} NT x NR x i=1 j=1 0014 00152 (5.4) where NT x , NRx , hi j are the number of transmit antennas, number of receive antennas and the corresponding channel coefficient. Figure 5.18(a) shows the eigenvalues distribution for the SPQHA antenna selection by maximising λ2 . The distribution of the first eigenvalue, λ1 is very similar in all cases. The distribution of λ2 is more of interest for comparison. The weakest λ2 is obtained when using 2x2 MIMO channel based on BSA−mid and BSB−mid which is approximately 24 dB lower than its λ1 at 99% probability level (Brown plot). The distribution of λ2 in this case will be used as reference. An improvement for λ2 of approximately 7 dB is obtained when using QHA-based MIMO at MIMOA1 (Green plot). Similar result can be seen with 4x2 MIMO channel from MIMOB1 (Blue plot). A further 4 dB improvement in λ2 distribution is obtained by using 8x2 MIMO channel from BSB (Purple plot). This is to be expected as it now has more antenna branches to improve diversity. By allowing spatial diversity in 8x2 MIMO channel formed with the cooperation between between BSA and BSB (MIMOA1 , MIMOB1 ), a further 5 dB gain is achieved (Red plot). This is a 16 dB improvement over the middle parasitic antenna MIMO (Brown plot). Finally, it can be seen in the figure that the distribution of λ2 of the MIMO antenna selection using all parasitic antennas (Black plot), is comparable to that of using 8x2 MIMO channel formed by MIMOA1 and MIMOB1 . Both setup are spatially separated and, it can be concluded that the diversity improvement of using QHA-based MIMO can be matched by the parasitic antenna selection. The distribution of the eigenvalue with the second parasitic antenna selection scheme which is to maximise λ1 is shown in Figure 5.18(b). Using this scheme, the λ1 for the MIMO SPQHA antenna (Black plot) is about 2 dB higher than that of 8x2 QHA-based MIMO with cooperative base station (Red plot). However, the λ2 for the MIMO SPQHA antenna selection is approximately 28 dB lower that its λ1 counterpart (at 99% probability level). Compared to the λ2 of the 8x2 QHA-based MIMO with cooperative base station, it is approximately 15 dB lower. Figure 5.19 shows the capacity comparison for four different MIMO antenna setup using equal power distribution. The first antenna setup uses the middle antenna, BA/B−mid from 124 5.11 Result III: Comparison with QHA-based MIMO Cumulative Probability 0 50 80 90 99 99.9 99.99 −50 −40 −30 −20 Magnitude, dB −10 0 10 0 10 (a) Maximising second eigenvalue, λ2 80 90 99 99.9 99.99 −50 −40 −30 −20 Magnitude, dB −10 (b) Maximising first eigenvalue, λ1 λ1: QHA MIMO 4x2 (MIMOA1) λ1: QHA 0 MIMO 4x2 (MIMOB1) 10 λ1: QHA MIMO 8x2 (MIMOB1, MIMOB2) λ : QHA MIMO 8x2 (MIMO , MIMO ) Cumulative Probability Cumulative Probability 0 50 1 A1 λ : SPQHA MIMO 2x2 (BS 1 B2 , BS A−mid ) Middle parasitic antennas B−mid −2 MIMO 2x2 (BS , BS ) All parasitic antennas λ1: SPQHA A B 10 λ2: QHA MIMO 4x2 (MIMOA1) λ2: QHA MIMO 4x2 (MIMOB1) λ2: QHA MIMO 8x2 (MIMOB1, MIMOB2) −4 λ : QHA 10 MIMO 8x2 (MIMO , MIMO ) 2 −40 A1 −20 B2 0 λ2: SPQHA MIMO 2x2 (BS , BSB−mid Magnitude, dB ) Middle parasitic antennas A−mid λ2: SPQHA MIMO 2x2 (BSA, BSB) All parasitic antennas (c) Legend Figure 5.18 MIMO eigenvalues distribution with two different schemes. 5.12 Result IV: Channel Characteristic 125 Figure 5.19 Capacity comparison for different MIMO schemes. each BS to form a 2x2 MIMO system. This setup is reprensented by the brown line and it will be called SPQHA MIMO 2x2 (Mid). The second setup utlitises MIMOA1 and MIMOB2 to form 8x2 MIMO system. This is represented by the red line and it is called QHA MIMO 8x2. The last two antenna scheme utilise MIMO antenna selection using the branches from the parasitic antennas on each base station. At one instance, only one parasitic antenna on each BS can be utilised to a 2x2 MIMO system. The antenna selection criteria is based on maximising λ2 (solid black line) and λ1 (dashed black line) which will be called SPQHA MIMO 2x2 (Max. λ2 ) and SPQHA MIMO 2x2 (Max. λ1 ), respectively. From the capacity figure, it can be seen SPQHA MIMO 2x2 (Mid) has the lowest capacity at 5.5 bits/s/Hz. This is to be expected it has the least degree of freedom. SPQHA MIMO 2x2 (Max. λ1 ) has better capacity at 6 bits/s/Hz. However, MIMO 2x2 (Max. λ1 ) has lower capacity than MIMO 2x2 (Max. λ2 ). In reference to Figure 5.18(b), even though MIMO 2x2 (Max. λ1 ) has higher first eigenvalue, its second eigenvalue is much lower which contribute less capacity. QHA MIMO 8x2 SPQHA and SPQHA MIMO 2x2 (Max. λ2 ) has similar capacity at approximately 6.5 bits/s/Hz. This again confirms that SPQHA with MIMO antenna selection with maximising second eigenvalue could match the performance of 8x2 MIMO QHA. 126 10 10 0 0 Magnitude, dB Magnitude, dB 5.12 Result IV: Channel Characteristic −10 −20 −30 −40 −50 5 Signal State(High/Low) 10 15 20 25 Relative time, s 30 −10 −20 −30 −40 −50 5 Signal State(High/Low) 10 (a) BSA−mid 15 20 25 Relative time, s 30 (b) BSB−mid Figure 5.20 A sample received signal at MSA1 from BSA−mid and BSB−mid (MeasFar−data ). 5.12 Result IV: Channel Characteristic A sample data of the received signal at MSA1 from BSA−mid and BSA−mid (MeasFar−data ) is shown in Figure 5.20. It can be seen that there are regions with high with low signal mean. The difference between the signal mean is around 15-20 dB. The high mean region occurs when there is a relatively clearer line of sight between the BS and MS. On the other hand, the signal mean attenuation, in the low region, is largely due to blockage by the users body with respect to the base station. It can also be observed that the signal variations in these two regions also varies significantly. In the low region, there are very large signal variations as compared to the signal in high region. The extraction of local mean and fast fading components has been discussed in Section 3.4.4. These two signal components will be analysed separately in the low and high region. Figure 5.21 shows the CDF plot for the fast fading signal component of the received signal at MSA1 from BSA−mid . Both signals exhibit Ricean distribution. The Ricean K-factor can be estimated using Eq. (3.8). The Ricean K-factor in the high and low state are 10.3 dB and 0.4 dB, respectively. Similar analysis can be performed for the signal from BSB−mid to MSA . The Ricean K-factor in the high and low state are 9.1 dB and 1.5 dB, respectively. This is very similar to that of BSA−mid . Figure 5.22 shows the CCDF plot for the local mean signal at MSA1 from BSA−mid . Both signals exhibit normal distribution. The distribution mean in the high and low state is 4.8 dB and −23.3 dB, respectively. Both state have similar standard deviation at 3.2 dB and 3.0 dB in the high and low region, respectively. Local mean distribution for the signal BSB−mid is 127 5.12 Result IV: Channel Characteristic 1 1 0.8 Cumulative probability Cumulative probability 0.9 0.7 0.6 0.5 0.4 0.3 0.2 0 0.2 0.4 0.6 0.8 1 Magnitude 1.2 1.4 0.6 0.4 0.2 High State Rician fit 0.1 0.8 Low State Rician Fit 0 1.6 0.5 (a) High State 1 1.5 Magnitude 2 2.5 (b) Low State Figure 5.21 Fast fading CDF plot for the received signal at MSA1 from BSA−mid (MeasFar−data ). Cumulative probability 1 0.8 0.6 0.4 High State Theoretical (High State) Low State Theoretical (Low State) 0.2 0 −20 −15 −10 Data −5 0 Figure 5.22 Local mean CDF plot for the received signal at MSA1 from BSA−mid (MeasFar−data ). 128 5.12 Result IV: Channel Characteristic S1−H,H Region High Low S3−H,L S2−L,L S3−L,H BSA−mid BSB−mid 0 5 10 15 Relative time, s 20 25 Figure 5.23 A sample of high and low regions for BSA−mid and BSB−mid with respect to MSA1 (MeasFar−data ). very similar. The mean in the high state is −4.0 dB with standard deviation of 3.3 dB. In the low state, the mean is −25.2 dB with standard deviation of 2.9 dB. Markov Chain Characteristic In the previous subsection, it has been shown that the link between the BSs and MS experiences high and low shadowing regions. Figure 5.23 shows the high and low regions over time for BSA−mid and BSB−mid with respect to MSA1 (MeasFar−data ). This shows that there is an opportunity to apply Markov chain approach to represent the high and low states. To represent dual base station setup, four state Markov chain can be used. Each state can be represented by [Sm−a,b ] where m is the state number, a represents the region for BSA and b represent the region for BSB . An example can be seen in Figure 5.23. The transition state probability, Pm,n is defined as Pm,n = Nm,n Nm (5.5) where Nm,n is the number of transition from Markov state m to state n and Nm is number in state m. The probability of each Markov state is represented by state probability matrix, W with element: Wn = Nn Nt (5.6) 129 5.13 Summary Table 5.6 Markov state and transition matrices. P 1 2 3 4 W 1 2 3 4 0.9895 0.0000 0.0288 0.0345 0.0000 0.9892 0.0096 0.0690 0.0026 0.0072 0.9615 0.0000 0.0053 0.0036 0.0000 0.8966 0.4798 0.3253 0.1313 0.0366 where Nt the total number. From MeasFar−data measurement data, these parameters are calculated and summarised in Table 5.6. However, since the data from this field measurement are not sufficient to calculated these parameters reliably, further field measurement campaign is needed and this will be left for future work. 5.13 Summary The performance and characteristics of the SPQHA in real environment has been successfully analysed. Antenna selection combining has been used to evaluated the diversity gain provided by the SPQHA. In this analysis, the antenna selection combining technique is divided into LAS, MAS and GAS. In LAS, the antenna selection is only performed on the local BS. With MAS, the antenna selection is performed between the middle parasitic antennas on both base station. Finally, GAS performs the antennas selection from all parasitic antennas on both base station. Local mean based diversity scheme with LAS has been shown to provide by up to 2 dB diversity gain. With MAS, it is able to achieve by up to 3 dB diversity gain. Finally, using GAS, 5 dB diversity could be achieved. With an instantaneous power diversity scheme, LAS provides 8 dB of diversity gain. With MAS, 8 dB of diversity gain could be achieved, which is due to spatial diversity. This indicates that SPQHA with LAS is able to provide pattern diversity. By combining spatial and pattern diversity, GAS is able to provide diversity by up to 13 dB. In this analysis, it was found that, the users location in the cell and mobile phone orientation has no significant impact on the diversity gain. SPQHA has also been shown to be able to reduce downlink interference. With LAS, even without any interference mitigation scheme, the SIR can be improved by approximately 7 dB. With MAS and GAS, SIR improvement in the order of 20 dB can be achieved. Using MIMO antenna selection, the performance of the MIMO SPQHA has been compared to the QHA-based MIMO. It was found the MIMO SPQHA antenna selection by maximising 5.13 Summary 130 the second eigenvalue is capable of matching the performance of the 8 element QHA-based MIMO. The channel characteristic analysis has also been performed. It was found that the channel exhibits Markov chain characteristic with high and low shadowing regions which correspond to the level of shadowing by the human body. There is a low region with 20 dB higher shadowing loss as compared to the high region. The high region also has a higher Ricean K-factor value which indicates the availability of a line-of-sight signal. The discussion above has presented the benefit of using SPQHA as analysed in the thesis. The SPQHA has three different states to provide radiation to left, middle and right. In a proper SPQHA, it will only has a single input and the radiation states are controlled by switches. This is beneficial as it reduces the number of RF-chains needed which can be expensive. The advantage of having the ability to change radiation pattern is that it could reduce the correlation between antenna branches. This in turn provide a better diversity gain. In the case of SPQHA, the diversity gain largely comes from pattern diversity as has been discussed. SPQHA with MIMO antenna selection has again showed the benefit of using SPQHA as compared to a full 8x2 QHA-based MIMO. Despite having similar performance, MIMO SPQHA only needs two RF-chains while QHA-based MIMO needs eight. Each RF-chain of the MIMO SPQHA has three states. Another advantage of having the ability to change the radiation pattern is that it could provide higher gain to the intended user or reduce gain to the unintended user. The improvement in the SIR valued as discussed confirms this. Chapter 6 Conclusion and Further Work 6.1 Conclusion As mentioned in the first chapter of this thesis, the main objectives of this thesis are to characterise the channel in public femtocell environment and to design suitable antennas for a public femtocell. These objectives have been successfully achieved and will be summarised in this section. Polarisation of the BS transmit antennas In Chapter 3, the performance of several BS transmit antennas with different polarisations against MSs has been investigated in anechoic chamber and field measurements. Both set of measurements have shown that neither VP nor HP base station antenna are optimum to provide the best performance in various scenario. From the anechoic chamber measurement, the performance different between VP and HP BS transmit antennas can be seen to be as high as 10 dB. The result from the field measurement campaign also shows similar trend. Even though the level of depolarisation by the large open space in the public femtocell environment is low, the polarisation mismatch between the BS and the MS originated from the random handling of the mobile stations by the users. The analysis have shown that a circularly polarised base station antenna is the best compromise and more effective in reducing polarisation mismatch loss due to human handlings compared to other polarisations under most scenarios. 6.1 Conclusion 132 QHA antenna design Two designs for QHA gain improvement have been successfully designed, fabricated and validated in Chapter 4. The first design is based on a parasitic meandered loop while the second design is based on a parasitic quadrifilar helix loop. Both designs are capable of producing up to 1.8 dB gain improvement in the boresight direction. Another parasitic element based QHA has also been designed and evaluated. By using a SPQHE at the side of the QHA, it gives the QHA the ability to tilt the beam to the side. The parasitic elements coupled the radiation from the radiating QHA and and subsequently re-radiated. The radiation from the SPQHE combines with the radiation from the driven QHA. The resulting radiation pattern is tilted away from the SPQHE. A separation distance of about 0.4λ has been found to be a good comprise in steered angle and return loss level. Even though smaller closer separation distance can tilt the beam more, it is at the expense of lower gain due to the increase in the return loss. Performance of the SPQHA antenna The performance and characteristics of the SPQHA in real environment have been successfully analysed in Chapter 5 . Antenna selection combining has been used to evaluate the diversity gain provided by the SPQHA. In this analysis, the antenna selection combining technique is divided into local antenna switching (LAS), Middle antenna switching (MAS) and global antenna switching (GAS). In LAS, the antenna selection is only performed on the local BS. With MAS, the antenna selection is performed between the middle parasitic antennas on both base stations. Finally, GAS performs the antenna selection from all parasitic antennas on both base stations. Local mean based diversity scheme with LAS has been shown to provide up to 2 dB diversity gain. With MAS, it is able to achieve up to 3 dB diversity gain. Finally, using GAS, 5 dB diversity could be achieved. With instantaneous power diversity scheme, LAS provides 8 dB of diversity gain. With MAS, 8 dB of diversity gain could be achieved, which is due to spatial diversity. This indicates that SPQHA with LAS is able to provide pattern diversity. By combining spatial and pattern diversity, GAS is able to provide diversity by up to 13 dB. In this analysis, it was found that, the users location in the cell and mobile phone orientation has no significant impact on the diversity gain. SPQHA has also been shown to be able to reduce co-channel interference. With LAS, even without any interference mitigation scheme, the SIR can be improved by approximately 7 dB. With MAS and GAS, SIR improvement in the order of 20 dB can be achieved. 6.2 Further work 133 Using MIMO antenna selection, the performance of the MIMO SPQHA has been compared to the QHA-based MIMO. It was found the MIMO SPQHA is capable of matching the performance of the 8 element QHA-based MIMO. The channel characteristic analysis has also been performed. It was found that the channel exhibits Markov chain characteristic with high and low shadowing regions. There is a low region with 20dB higher shadowing loss as compared to the high region. The high region also has a higher Ricean K-factor value which indicates the availability of a line-ofsight signal. 6.2 6.2.1 Further work Further design improvement for the Side Parasitic QHA A further design and analysis is required to improve and verify the performance of the SPQHA. In this work, the SPQHA was fabricated as a separate unit to represent each parasitic radiation pattern. In real SPQHA, only one radiating antenna will be needed and the side parasitic elements will be controlled by switches. The effect of these switched has not been evaluate in this work. A reactive-loaded SPQHA has not been full evaluated yet through this work. With reactive loading, the SPQHA shall be able to continuous steering the beam instead of discrete beamsteering as in current design. 6.2.2 Nested Parasitic Quadrifilar Helix Loop (PQHL) In this work, the PQHL has been shown to be able to improve the gain of the QHA. Even though PQHL does not increase the radius of the antenna setup, it increases the axial length. There is an opportunity to nest the PQHL into the QHA. However, preliminary work has shown that nested PQHL reduces the input impedance and bandwidth. Despite this drawback, nested PQHL does not affect the volume of the QHA at all. Further work are need to improve the performance of the PQHL in nested configuration. 6.2.3 QHA Feed Network Improvement The quadrature feed network used through the analysis of this work has some drawbacks. The size of the feed network is rather large. Since the phase shifting depends on the microstrip length, a substrate with higher permittivity will be able to shrink the feed network to a certain degree. Another issue with quadrature feed network is that interfere with operation 6.2 Further work 134 of the QHA as compared to when the QHA is in free space. A new feed network design is required to alleviate these issues. 6.2.4 Further Field Measurement for Markov Chain Parameter It has been identified that the signal ink between the BS and MS exhibit Markov chain characteristic. The current measurement data is not enough to extract the Markov chain parameter. Another field measurement will be need to get a reliable Markov chain parameters. 6.2.5 Evaluation for Large Scale of SPQHA In this work, each base station has only been equipped with one set of SPQHA. The results have shown that the MIMO antenna selection based on the SPQHA is capable of providing similar diversity benefit compared to a QHA-base MIMO. It would be beneficial to evaluate the performance of an large scale SPQHA consisting on multiple SPQHAs on each BS and compare its performance with MIMO antennas. There is a high potential to achieve a standard massive MIMO benefit using MIMO antenna selection based on large scale SPQHA deployment and further evaluations are needed. 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Appendix A Derivation of Polarisation Efficiency for transmission using LP and CP antennas When a LP signal is received by CP antenna (or vice versa), polarisation mismatch loss will occur. The polarisation unit vector for LP with arbitrary slant angle θ and RHCP are given by Eq. (2.2) and Eq. (2.3). From Eq. (2.1), the polarisation efficiency, PE for these antennas combination can be calculated as follows ∗ 2 | PE = |ρLP • ρRH 0015∗ 2 0014 1 ˆ = [cos θ xˆ + sin θ y] ˆ • √ (xˆ − jy) 2 2 1 j = √ cos θ + √ sin θ 2 2 0001 1 = cos2 θ + sin2 θ 2 (A.1) (A.2) (A.3) (A.4) From trigonometric identity, cos2 θ +sin2 θ is equals to 1 irrespective of slant angle θ . From this result, PE can be simplified as follows PE = 1 2 (A.5) |
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